Driving circuit for driving transistor based on control current

ABSTRACT

A switching circuit includes: a transistor having a first electrode, a second electrode and a control electrode; a zener diode; and a capacitor. A connection between the first electrode and the second electrode is capable of temporally switching between a conduction state and a non-conduction state by switching a control voltage of the transistor. The zener diode and the capacitor are coupled in series between the first electrode and the control electrode of the transistor. The first electrode is a drain or a collector.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.13/105,021 filed on May 11, 2011 now issued as U.S. Pat. No. 8,179,169and entitled DRIVING CIRCUIT WITH VARIABLE RESISTOR FOR TRANSISTOR,which is a divisional of U.S. patent application Ser. No. 12/654,323filed on Dec. 17, 2009, now issued as U.S. Pat. No. 7,982,508 andentitled SWITCHING CIRCUIT AND DRIVING CIRCUIT FOR TRANSISTOR, which isa divisional U.S. patent application Ser. No. 11/723,967 filed on Mar.22, 2007, now issued as U.S. Pat. No. 7,671,636 and entitled SWITCHINGCIRCUIT AND DRIVING CIRCUIT FOR TRANSISTOR, and is based on JapanesePatent Applications No. 2006-79382 filed on Mar. 22, 2006, No.2006-86225 filed on Mar. 27, 2006, No. 2006-204769 filed on Jul. 27,2006, No. 2006-204770 filed on Jul. 27, 2006, No. 2006-243832 filed onSep. 8, 2006, No. 2007-73381 filed on Mar. 20, 2007, No. 2007-73382filed on Mar. 20, 2007, and No. 2007-74030 filed on Mar. 22, 2007, thedisclosures of which are incorporated herein by reference.

FIELD OF THE INVENTION

The present invention relates to a switching circuit and a drivingcircuit for a transistor.

BACKGROUND OF THE INVENTION

Switching circuits have been widely utilized in DC-DC converters,inverters, or the like. There are many possibilities that typicalswitching circuits utilize transistors such as MOSFETs, IGBTs, or thelike. In switching circuits, since voltages applied to gates oftransistors are switched, signal paths between major electrodes (namely,sources and drains in case of unipolar transistors such as MOSFET etc.,collectors and emitters in case of bipolar transistors such as IGBTetc.) of the transistors are switched in a temporal manner between aconducting state and a non-conducting state.

Various sorts of requirements have been made with respect to switchingcircuits, namely, these switching circuits are required to be madecompact, and are required to be arranged by employing a small number ofcircuit components; ON-resistances of the switching circuits must below, and switching losses thereof must be low; and surge voltages of theswitching circuits must be suppressed to low surge voltages.

In order to make the switching circuits compact, there is a merit iffrequencies at which gate voltages are changed are increased. Thisreason is given that since the frequencies are made high, inductancesand capacitances required by the switching circuits can be suppressed tosmall values.

If operating frequencies of switching circuits are increased, thenswitching losses of transistors may cause serious problems. In order tosuppress these switching losses to small values, there is a merit ifswitching speeds of the transistors are increased. Sinceresearching/developing operations of these transistors are progressed,the switching speeds thereof are increased.

However, even in the present stage, it is practically difficult tosuppress both the switching losses and the surge voltages at the sametime. In order to suppress the switching losses to small values, it isadvantage to increase the switching speeds of the transistors. However,if the switching speeds are increased, then the surge voltages areincreased. In order to suppress the surge voltages to low voltages, itis advantage to decrease the switching speeds of the transistors.However, if the switching speeds are decreased, then the switchinglosses are increased. A trade-off relationship is present between theswitching losses and the surge voltages. If the switching losses aresuppressed to the small values, then the surge voltages are increased,whereas if the surge voltages are suppressed to the low voltages, thenthe switching losses are increased. It is practically difficult thatboth the switching losses and the surge voltages are suppressed to thelow values at the same time.

In order to suppress both a surge voltage and a switching loss to a lowsurge voltage and a low switching loss, such a gate driving circuit forswitching gate resistors in a stepwise manner in connection with a timeelapse has been proposed. However, this technical idea has problems.That is, the gate driving circuit becomes complex, and a total number ofelectronic components required to manufacture a switching circuit isnecessarily increased.

Under such a circumstance, the transistor described in JP-A-2004-6598(corresponding to U.S. Pat. No. 6,700,156) has been developed. Thistransistor is a MOSFET in which a p layer containing a p type impurityin low concentration has been formed at a position located adjusted to ap base layer.

In the case of a normal MOSFET, the higher a voltage between the sourceand the drain of the normal MOSFET becomes, the lower a capacitancebetween the drain and the gate thereof is decreased. When thecapacitance between the drain and the gate is small, a voltage changespeed between the drain and the source becomes high, which is generatedwhen the transistor is turned OFF. In the normal MOSFET, the higher thevoltage between the drain and the source is increased when the MOSFET isturned OFF, the higher the voltage change speed between the drain andthe source becomes, so that a surge voltage is increased.

As described in JP-A-2004-6598, if the p layer containing the p typeimpurity in the low concentration is added to the position locatedadjacent to the p base layer, then the following characteristic can beobtained. That is, the higher the voltage between the drain and thesource becomes, the larger the capacitance between the gate and thedrain is increased. When this MOSFET is employed, if the transistor isturned OFF so that the voltage between the drain and the source isincreased, then the capacitance between the gate and the drain isincreased. As a result, the voltage change speed between the drain andthe source is decreased, so that the surge voltage can be suppressed tothe low surge voltage.

However, in the above-described MOSFET, the below-mentioned problemoccurs. That is, the current flows through the p layer containing the ptype impurity in the low concentration, so that the ON resistancethereof is increased. Also, the stronger the surge voltage issuppressed, the higher the ON resistance of the transistor is increased.Furthermore, in order to effectively suppress the surge voltage, the player containing the p type impurity in the low concentration must bemade large. If the p layer is made large, then the cell pitch of thetransistor becomes large, so that the ON resistance of the transistor isincreased. In other words, in accordance with the measure manner forimproving the internal structure of the transistor, it is practicallydifficult to suppress the ON resistance, the surge voltage, and theswitching loss to the lower values.

As a consequence, it is desirable to accomplish switching circuitscapable of suppressing ON resistances to lower ON resistance values,capable of suppressing switching losses to lower switching losses, andalso capable of suppressing surge voltages to lower surge voltages,while the switching circuits can be constructed by employing smallnumbers of structural components.

Also, other switching circuits are known in the field. In the switchingcircuits, power supplies and loads are series-connected between one pairof major electrodes of transistors; and since the transistors are turnedON/OFF, conditions under which electric power is supplied to the loads,and conditions under which the electric power is not supplied areselectively switched. For instance, while an inverter circuit contains aswitching circuit, a transistor of the switching circuit is turnedON/OFF in order that DC power is converted into AC power, and then, theconverted AC power is supplied to a motor (one example of load).

One example as to this sort of switching circuits has been disclosed inJP-A-6-326579. The switching circuit has been equipped with a seriescircuit connected between a gate electrode and a drain electrode of atransistor, in which a zener diode has been series-connected to a diode.The zener diode has been set in such a manner that when a surge voltageis generated on a wiring line on the drain electrode side of thetransistor, the zener diode may break down in response to the generatedsurge voltage. The function of the diode is to avoid that a gate currentflowing to the gate electrode of the transistor when the transistor isturned ON flows via the series circuit to the wiring line on the drainelectrode side of the transistor.

In the above-described switching circuit, when such a surge voltage isgenerated on the wiring line on the drain electrode side of thetransistor, the zener diode breaks down. This surge voltage is higherthan a total voltage (V_(ZD)+V_(F)) of a breakdown voltage “V_(ZD)” ofthe zener voltage and a forward direction voltage “V_(F)” of the diode.When the zener diode breaks down, the current flows from the wiring lineof the drain electrode side of the transistor via the series circuit tothe gate electrode of the transistor, so that the voltage of the gateelectrode is increased, and thus, the transistor is turned ON. Thisswitching circuit can be operated in such a manner that when the surgevoltage is generated, since the transistor is instantaneously turned ON,the surge energy may be discharged via the transistor to the externalcircuit.

However, the breakdown voltage V_(ZD) of the zener diode containsfluctuations of approximately of ±10% due to manufacturing tolerance. Asa consequence, in this switching circuit, depending upon zener diodes tobe employed, timing when these zener diodes break down is fluctuated. Asa result, the above-described switching circuit has such a problem thatthe timing at which the surge voltage is discharged outside the ownswitching circuit cannot be set in higher precision.

Another type of switching circuit is desirable. That is, not only timingfor slowing a changing speed of a voltage at a control electrode of atransistor is made stable, but also the changing speed of the voltage atthe control electrode of the transistor is slowed at earlier timing thansuch a timing when a voltage at a major electrode of the transistor onthe high voltage side exceeds a power supply voltage, so that thesuppression capability for the surge voltage can be improved.

Also, in a switching circuit equipped with a transistor (MOSFET), anadverse influence caused by a surge voltage in the case that the MOSFETis brought into a turn-OFF state is required to be reduced.

As indicated in FIG. 34, a load driving circuit capable of protecting aMOSFET 5 from a surge voltage by the circuit itself is disclosed inJP-A-6-326579. A diode 9 a and a zener diode 9 have beenseries-connected to a branch line 310 under such a condition that thediode 9 a is located opposite to the zener diode 9. The diode 9 a playsa role of suppressing that a current flows through the branch line 310when the MOSFET 5 is turned ON in response to an ON signal outputtedfrom an output terminal 318 of a control circuit 3. The zener diode 9plays a role of increasing a voltage at a gate G of the MOSFET 5 in sucha case that a positive surge voltage is applied to a power supply line322 when the MOSFET 5 is turned OFF. The positive surge voltage ishigher than, or equal to the zener voltage “V_(ZD)” plus the diodeforward direction threshold voltage “V_(F).” As a result, the MOSFET 5is brought into an ON state, so that a signal path between the sourceand the drain of the MOSFET 5 becomes conductive, and thus, the surgevoltage can be discharged.

However, normally, a zener voltage “V_(ZD)” of a zener diode isfluctuated by approximately ±10%. Accordingly, in such a case that thezener diode is employed as explained in the above-described conventionaltechnique, it is practically difficult that the reference voltage(V_(ZD)+V_(F)) for suppressing the serge voltage is correctly set to apredetermined value.

FIG. 35 represents a result obtained by simulating a temporal change ofa drain voltage VD when a MOSFET is turned OFF in such a case that avoltage of a power supply PS is set to 100V. FIG. 35 shows a simulationresult in the case that no surge voltage suppression measure isperformed in the switching circuit (indicated by line “XXXVA”); asimulation result in the case that the serve voltage suppression measureusing the zener diode 9 is performed as explained in the above-describedconventional load driving circuit, and then, “V_(ZD)+V_(F)” is set to100 V (represented by line “XXXVB”); and a simulation result in the casethat although the serve voltage suppression measure using the zenerdiode 9 is performed as explained in the above-described conventionalload driving circuit, “V_(ZD)+V_(F)” is varied to 90 V and 110 V due tofluctuations of the zener diode “V_(ZD)” (indicated by lines “XXXVC” and“XXXVD”).

Assuming now that a surge voltage is defined as a difference from a peakvalue of the drain voltage VD up to a value under steady condition, whenthe surge voltage suppression measure is not taken, the surge voltage is22V; and when “V_(ZD)+V_(F)” is set to 100 V, the surge voltage is 3 V,so that the suppression effect of the surge voltage may become achieved.However, if (V_(ZD)+V_(F)) becomes 110 V, then the surge voltage becomes13 V, so that the suppression effect of the surge voltage isconsiderably lowered. Also, when (V_(ZD)+V_(F)) becomes 90 V, the drainvoltage VD cannot be recovered to 100 V corresponding to the powersupply voltage.

As previously described, in the surge voltage suppressing circuits usingthe zener diodes, when V_(ZD)+V_(F) can be correctly controlled to thepredetermined value, the surge voltage suppressing effect can beachieved. However, as explained above, since the zener voltage V_(ZD) isfluctuated, the suppression effect of the surge voltage may not besufficiently achieved, or the drain voltage VD may not be recovered upto the power supply voltage.

As a consequence, such a switching circuit capable of suppressing asurge voltage under stable condition is required, and another switchingcircuit capable of reducing both a switching loss and noise of atransistor is required.

Also, another switching circuit is known in the field. In the switchingcircuit, since a transistor connected to a load is turned ON/OFF, acondition under which electric power is supplied to the load, andanother condition under which the electric power is not supplied areselectively switched. For instance, in an inverter circuit, a transistoris turned ON/OFF in order that DC power is converted into AC power, andthen, the converted AC power is supplied to a motor. The turn-ON/OFFoperations of the transistor of this sort of circuit are controlled by adriving circuit connected to the gate electrode (otherwise, baseelectrode) of this transistor.

FIG. 65A to FIG. 65E show an example of operating waveform diagrams asthe related art in such a case that a field-effect type transistor isemployed as this sort of transistor. A driving circuit switches ON/OFFoperations of the field-effect transistor by applying a driving voltage“Vin” to a gate electrode of the field-effect transistor.

A first description is made of a transition time period during which thefield-effect transistor is turned ON. When the driving voltage Vinbecomes a high level from a low level, a positive gate current “Ig”flows toward the gate electrode of the transistor, so that electroncharges are stored in the gate electrode. When the electron charges arestored in the gate electrode, a gate-to-source voltage “Vgs” of thetransistor is increased. When the gate-to-source voltage “Vgs” of thetransistor is increased, a drain current “Id” starts to flow from thedrain of the transistor to the source thereof, so that a drain-to-sourcevoltage Vds is decreased. The turn-OFF operation of the transistor istransferred to the turn-ON operation via these operation stages.

Next, a description is made of a transition time period T100 duringwhich the field-effect transistor is turned OFF. When the drivingvoltage Vin becomes a low level from a high level, the electron chargesstored in the gate electrode are discharged, and a negative gate current“Ig” flows from the gate electrode toward the driving circuit, so thatthe gate-to-source voltage “Vgs” of the transistor is decreased. Whenthe gate-to-source voltage “Vgs” of the transistor is decreased, thedrain current “Id” is also decreased, so that the drain-to-sourcevoltage Vds is increased. The turn-ON operation of the transistor istransferred to the turn-OFF operation via these operation stages.

As shown in FIG. 65A to FIG. 65E, in a final stage of the transitiontime period T100 during which the transistor is turned OFF, a surgevoltage is generated in the drain-to-source voltage Vds. This surgevoltage is induced by the drain current Id which is steeply varied, andan inductance staying on the wiring line of the drain electrode sidewithin the circuit.

In order to suppress the increase of this surge voltage, the draincurrent Id may be gently varied. For instance, if the gate resistance ofthe transistor is increased, then the speed at which the electroncharges stored in the gate electrode are discharged is decreased, sothat the negative gate current Ig flows in a gentle manner. As a result,the drain current Id is also gently decreased, so that increasing of thesurge voltage can be suppressed. However, if the drain current Id of thetransistor is gently decreased, then a time period required until thetransistor is turned OFF is increased, so that a turn-OFF loss isincreased. That is to say, this sort of transistors have a trade-offrelationship between the surge voltage and the turn-OFF loss during thetransition time period T100 of the turn-OFF operation of the transistor.

In order to overcome this trade-off relationship, it is desirable tosteeply vary the drain current Id in the beginning stage of thetransition time period T100 of the turn-OFF operation, and also, it isdesirable to gently vary the drain current Id in the final stage of thetransition time period T100 of the turn-OFF operation. If the draincurrent Id is steeply varied in the beginning stage of the transitiontime period T100, then the time required for turning OFF the transistorcan be shortened. As a result, the turn-OFF loss can be suppressed to asmall loss. If the drain current Id is gently varied in the final stageof the transition time period T100, then increasing of the surge voltagecan be suppressed.

JP-A-6-291631 discloses such a technical idea capable of adjusting aresistance value of a gate resistor of a transistor based upon voltagesbetween major electrodes of the transistor. As the voltages between themajor electrodes, there are a voltage between a drain electrode and asource electrode, a voltage between a collector electrode and an emitterelectrode, and so on. In this technical idea, the following adjustmentis carried out: That is, when the voltage between the major electrodesof the transistor is high, the resistance value of the gate isincreased, whereas when the voltage between the major electrodes of thetransistor is low, the resistance value of the gate is decreased.Concretely speaking, this driving circuit has been equipped with aresistance variable means connected to the gate electrode of thetransistor. The resistance variable means has been arranged by asemiconductor switching element and a fixed resistor which isparallel-connected to the semiconductor switching element. When thevoltage between the major electrodes of the transistor is higher than apredetermined value, the semiconductor switching element is turned OFF,whereas when the voltage between the major electrodes of the transistoris lower than the predetermined value, the semiconductor switchingelement is turned ON. In other words, when the voltage between the majorelectrodes of the transistor is high, the semiconductor switchingelement is turned OFF, so that the gate resistance is adjusted to beincreased in response to a resistance value of a fixed resistor. Whenthe voltage between the major electrodes of the transistor is low, thesemiconductor switching element is turned ON, so that the gateresistance is adjusted to be decreased in response to an internalresistance value of the semiconductor switching element.

When the above-described driving circuit is utilized, in the beginningstage (when voltage between major electrodes is low) of the transitiontime period for the turning-OFF operation of the semiconductor switchingelement, the semiconductor switching element is turned ON, so that theresistance value of the gate resistor is adjusted to become small, andthus, the gate current is steeply varied. As a result, the drain currentof the transistor is steeply varied, so that the time required forturning OFF the semiconductor switching element can be shortened.Furthermore, in the final stage (when voltage between major electrodesis high) of the transition time period for the turning-OFF operation,the semiconductor switching element is turned OFF, so that theresistance value of the gate resistor is adjusted to become large, andthus, the gate current is gently varied. As a result, the drain currentof the transistor is gently varied, so that increasing of surge voltagecan be suppressed.

As a consequence, in the above-described driving circuit, the highresistance state of the gate resistor is realized by utilizing the fixedresistor having the high resistance value. In order to suppressincreasing of the surge voltage, it is desirable that the resistancevalue of the fixed resistor is set to be large. However, the fixedresistor having the higher resistance value may increase the turn-OFFloss. As a consequence, in order to suppress increasing of the turn-OFFloss, such a timing when the resistor is switched to the fixed resistorhaving the high resistance value by turning ON/OFF the semiconductorswitching element must be correctly set to the final stage of thetransition time period for turning ON/OFF the semiconductor switchelement. In the final stage of the transition time period for turningON/OFF the semiconductor switching element, the voltage between themajor electrodes of the transistor has been reached to the highervoltage state. In the above-described driving circuit, the turn ON/OFFoperations of the semiconductor switching element must be controlled bycorrectly changing the voltage between the major electrodes of thistransistor up to the threshold value as to the turn-ON/OFF operations ofthe semiconductor switching element. As a consequence, in order torealize such a circuit, a total number of necessary circuit componentsis increased, so that cost of this circuit is necessarily increased.

Another technical idea is required which may adjust the resistance valueof the gate resistor of the transistor based upon the voltage betweenthe major electrodes of the transistor in accordance with another mannerdifferent from the above-described manner. It should also be noted thatthe problems have been described by mainly considering the transitiontime period for turning OFF the transistor in the above-mentioneddescriptions. However, even in a transition time period for turning ONthe transistor, there are many possibilities that the resistance valueof the gate resistor of the transistor is wanted to be adjusted basedupon the voltage between the major electrodes of the transistor. Inother words, such a technical idea capable of achieving useful resultsis required even in any transition time periods for turning ON andturning OFF the transistor.

Also, JP-A-1-183214 discloses a circuit for diving a bipolar typetransistor. It should also be noted that the technical idea related tothis driving circuit may also be utilized in such a case that afield-effect type transistor is driven.

The above-described driving circuit has been equipped with two resistorsconnected to a gate electrode of a unipolar type transistor. Inaccordance with this driving circuit, in a beginning stage of atransition time period for turning OFF the transistor, a negative gatecurrent from the gate electrode flows through the two resistors. On theother hand, in a final stage of the transition time period for turningOFF the transistor, a negative gate current from the gate electrodeflows only through the other resistor, while one resistor is cut off.

If the above-described driving circuit is utilized, then the negativegate current is steeply varied in the beginning stage of the transitiontime period for turning OFF the transistor, so that the drain current issteeply varied, and thus, the time required for turning OFF thetransistor can be shortened. Moreover, if the above-described drivingcircuit is utilized, then the negative gate current is gently varied inthe final stage of the transition time period for turning OFF thetransistor, so that the drain current is gently varied, and thus,increasing of the surge voltage can be suppressed.

In the above-described driving circuit, timing for cutting off oneresistor has been previously set based upon a time constant as to both acapacitor and a resistor assembled in the driving circuit. As aconsequence, when the turn-OFF operation is repeatedly carried out, suchan event may occur. That is, the timing for cutting off one resistor isdeviated from such a timing for determining both the beginning stage andthe final stage of the turn-OFF operation. The manner for controllingthe cut-off timing which has been previously set cannot be synchronizedwith the operation of the transistor. As a consequence, increasing ofthe surge voltage and increasing of the turn-OFF loss cannot be firmlysuppressed.

As a consequence, the below-mentioned technical idea is desired. Thatis, while monitoring such a condition that a transistor is operated, aresistance value of a gate resistor of the transistor is adjusted. Itshould also be understood that the problems have been described bymainly considering the transition time period for turning OFF thetransistor in the above-mentioned descriptions. However, even in atransition time period for turning ON the transistor, there are manypossibilities that the below-mentioned technical idea is required. Thatis, while monitoring such a condition that the transistor is operated,the resistance value of the gate resistor of the transistor is adjusted,even in the transition time period for turning ON the transistor. As aconsequence, such a technical idea capable of achieving useful resultsis required even in any of the transition time periods for turning OFFand turning ON the transistor.

SUMMARY OF THE INVENTION

In view of the above-described problem, it is an object of the presentinvention to provide a switching circuit having a small switching lossand a small surge voltage. It is another object of the present inventionto provide a driving circuit for a transistor.

According to a first aspect of the present disclosure, a switchingcircuit includes: a transistor having a first electrode, a secondelectrode and a control electrode; a zener diode; and a capacitor. Aconnection between the first electrode and the second electrode iscapable of temporally switching between a conduction state and anon-conduction state by switching a control voltage of the transistor.The zener diode and the capacitor are coupled in series between thefirst electrode and the control electrode of the transistor. The firstelectrode is a drain or a collector.

In the above switching circuit, the switching loss can be reduced.Further, the surge voltage also can be reduced. Accordingly, both of theswitching loss and the surge voltage are limited to be small.Furthermore, the transistor has a low on-state resistance (or, a lowon-state voltage), and the number of parts for providing the circuit issmall.

According to a second aspect of the present disclosure, a switchingcircuit includes: a transistor having a first electrode, a secondelectrode and a control electrode; a zener diode; and a capacitor. Aconnection between the first electrode and the second electrode iscapable of temporally switching between a conduction state and anon-conduction state by switching a control electrode voltage of thetransistor. The zener diode and the capacitor are coupled in seriesbetween the first electrode and the second electrode of the transistor.The first electrode is a drain or a collector. The second electrode is asource or an emitter.

In the above switching circuit, both of the switching loss and the surgevoltage are limited to be small. Furthermore, the transistor has a lowon-state resistance (or, a low on-state voltage), and the number ofparts for providing the circuit is small.

According to a third aspect of the present disclosure, a switchingcircuit includes: a transistor having a high voltage side mainelectrode, a low voltage side main electrode and a control electrode,wherein a power source and a load are coupled in series between the highvoltage side main electrode and the low voltage side main electrode; acontrol circuit for outputting a voltage, which provides to switch thetransistor on and off, wherein the control circuit is coupled with thecontrol electrode of the transistor; a series circuit having a firstcapacitor and a first diode, wherein the series circuit is coupledbetween the control electrode and the high voltage side main electrodeof the transistor, wherein the first capacitor and the first diode arecoupled in series with each other, wherein a cathode of the first diodeis coupled with a control electrode side, and wherein an anode of thefirst diode is coupled with a high voltage side main electrode side; anda voltage control circuit. The voltage control circuit is coupled with aconnection portion between the first capacitor and the first diode. Thevoltage control circuit controls a voltage of the connection portion.

In the above circuit, by utilizing the voltage control circuit, a timingfor slowing a changing speed of a voltage generated in the controlelectrode of the transistor is controlled when the transistor is turnedoff. When this timing is adjusted earlier, the surge voltage can bereduced. Further, in the above circuit, since the changing speed of thevoltage generated in the control electrode of the transistor is slowedduring the transistor is turning off, both of the turn-off loss and thesurge voltage are reduced.

According to a fourth aspect of the present disclosure, a switchingcircuit includes: a transistor having a first electrode, a secondelectrode and a control electrode, wherein the transistor controls afirst electrode current flowing through the first electrode inaccordance with a signal inputted in the control electrode; a capacitor;and a diode having an anode terminal and a cathode terminal. The firstelectrode of the transistor is coupled with the anode terminal of thediode through the capacitor therebetween. The control electrode of thetransistor is coupled with the cathode terminal of the diode. Thecontrol electrode is a gate or a base. The first electrode is a drain ora collector.

In the above circuit, the surge voltage of the switching circuitincluding the transistor is accurately and stably reduced.

According to a fifth aspect of the present disclosure, a switchingcircuit includes: a transistor having a first electrode, a secondelectrode and a control electrode, wherein the transistor controls afirst electrode current flowing through the first electrode inaccordance with a signal inputted in the control electrode; a capacitor;and a diode having an anode terminal and a cathode terminal. The firstelectrode of the transistor is coupled with the anode terminal of thediode. The control electrode of the transistor is coupled with thecathode terminal of the diode through the capacitor. The controlelectrode is a gate or a base. The first electrode is a drain or acollector.

In the above circuit, the surge voltage of the switching circuitincluding the transistor is accurately and stably reduced.

According to a sixth aspect of the present disclosure, a driving circuitfor driving a transistor includes a variable resistor. The transistorincludes a control electrode, a first electrode and a second electrode.The variable resistor is coupled with the control electrode of thetransistor. The variable resistor includes a depletion layer, which isexpandable in accordance with a voltage between the first electrode andthe second electrode of the transistor. The depletion layer is capableof controlling a current path of the variable resistor.

In the above driving circuit, there is no need to have a circuit forsetting a threshold accurately, similar to on/off operation of asemiconductor switching element. Accordingly, the construction of theabove driving circuit is simplified. In the above driving circuit, awidth of the current path of the variable resistor is controlled byutilizing the depletion layer, which is expandable in accordance withthe voltage between the main electrodes of the transistor. Further, theconstruction of the circuit is simplified, and a manufacturing cost ofthe circuit is also reduced.

According to a seventh aspect of the present disclosure, a drivingcircuit for driving a transistor includes a control circuit. Thetransistor includes a control electrode, an output electrode and aninput electrode, and the control circuit controls a control electroderesistance of the transistor based on a control current flowing throughthe control electrode of the transistor.

In the above driving circuit, the resistance of the gate resistor in thetransistor is controlled based on the gate current sufficientlycorresponding to the state of operation of the transistor. Accordingly,the above driving circuit can control the resistance of the gateresistor in synchronous with the operation of the transistor.

BRIEF DESCRIPTION OF THE DRAWINGS

The above and other objects, features and advantages of the presentinvention will become more apparent from the following detaileddescription made with reference to the accompanying drawings. In thedrawings:

FIG. 1 is a circuit diagram for showing a switching circuit of a firstembodiment of the present invention;

FIG. 2 is a graph for representing temporal changes of drain voltages,which occur before and after when transistors of the first embodimentand of a comparison example are turned OFF;

FIG. 3 is a graph for indicating temporal changes in a drain current anda drain voltages, which occur before and after when the transistor isturned OFF under such a condition that a signal path between a drain anda gate of the transistor are opened;

FIG. 4 is a graph for representing temporal changes of a drain currentand a drain voltage, which occur before and after the transistor isturned OFF under such a condition that the signal path between the drainand the gate is connected by a series circuit constructed of a zenerdiode and a capacitor;

FIG. 5 is a graph for representing temporal changes of a drain currentand a drain voltage, which occur before and after the transistor isturned OFF under such a condition that the signal path between the drainand the gate is connected by the series circuit constructed of the zenerdiode and the capacitor, while a zener voltage of the zener diode isdifferent from that of FIG. 4;

FIG. 6 is a graph for showing a relationship between switching lossesand surge voltages of the first embodiment and of the comparisonexample;

FIG. 7 is a circuit diagram for showing a switching circuit of a secondembodiment of the present invention;

FIG. 8 is a graph for representing temporal changes of drain voltages,which occur before and after when transistors of the second embodimentand of the comparison example are turned OFF;

FIG. 9 is a graph for representing temporal changes of a drain currentand a drain voltage, which occur before and after the transistor isturned OFF under such a condition that a signal path between a drain anda gate is connected by a series circuit constructed of a zener diode anda capacitor;

FIG. 10 is a graph for representing temporal changes of a drain currentand a drain voltage, which occur before and after the transistor isturned OFF under such a condition that the path between the drain andthe gate is connected by the series circuit constructed of the zenerdiode and the capacitor, while a zener voltage of the zener diode isdifferent from that of FIG. 9;

FIG. 11 is a graph for showing a relationship between switching lossesand surge voltages of the second embodiment and of the comparisonexample;

FIG. 12A is a sectional view for indicating a semiconductor substratewhere a transistor, a zener diode, and a capacitor of a switchingcircuit of a third embodiment of the present invention have been formed,and FIG. 12B is a circuit diagram for indicating an equivalent circuitof the structure of FIG. 12A;

FIG. 13 is a sectional view for indicating a semiconductor substratewhere a transistor, a zener diode, and a capacitor of a switchingcircuit according to one modification of the third embodiment have beenformed;

FIG. 14 is a sectional view for indicating a semiconductor substratewhere a transistor, a zener diode, and a capacitor of a switchingcircuit according to another modification of the third embodiment havebeen formed;

FIG. 15 is a sectional view for indicating a semiconductor substratewhere a transistor, a zener diode, and a capacitor of a switchingcircuit according to another modification of the third embodiment havebeen formed;

FIG. 16A is a sectional view for indicating a semiconductor substratewhere a transistor, a zener diode, and a capacitor of a switchingcircuit according to a further modification of the third embodiment havebeen formed, and FIG. 16B is a circuit diagram for showing an equivalentcircuit of the structure of FIG. 16A;

FIG. 17 shows a basic circuit diagram of a switching circuit of a fourthembodiment of the present invention;

FIG. 18A is a graph for representing a variation of a voltage of a drainelectrode of a transistor employed in a switching circuit where a surgevoltage measure circuit is not provided, FIG. 18B is a graph forrepresenting a variation of a voltage of a drain electrode of atransistor employed in a switching circuit where only a series-circuitwithin the surge voltage measure circuit is provided, and FIG. 18C is agraph for showing a variation of a voltage of a drain electrode of atransistor employed in the switching circuit of the fourth embodiment;

FIG. 19 is a circuit diagram for showing a switching circuit of thefourth embodiment;

FIG. 20A is a graph for indicating time elapse changes in a voltage VGof a gate electrode of the transistor, a voltage VD of the drainelectrode D, and voltages V1 and V2 in the switching circuit of thefourth embodiment; and FIG. 20B is a graph for showing time elapsechanges in a drain current ID of the transistor, and currents I1 and I2in the switching circuit of the fourth embodiment;

FIG. 21 is a graph for showing a variation of a voltage of the drainelectrode in a transition period when the transistor is turned OFF;

FIG. 22 is a graph for showing a variation of a drain current in thetransition period when the transistor is turned OFF;

FIG. 23 is a graph for indicating a relationship between a turn-OFF lossand a surge voltage of the switching circuit of the fourth embodiment;

FIG. 24A is a graph for showing time elapse changes in a voltage VG of agate electrode, a voltage VD of a drain electrode D of a transistor, andvoltages V1 and V2 in a switching circuit of a comparison example 1; andFIG. 24B is a graph for indicating time elapse changes in a draincurrent ID of a transistor, and currents I1 and I2 in the switchingcircuit of the comparison example 1;

FIG. 25A is a graph for showing time elapse changes in a voltage VG of agate electrode, a voltage VD of a drain electrode D of a transistor, andvoltages V1 and V2 in a switching circuit of a comparison example 2; andFIG. 25B is a graph for indicating time elapse changes in a draincurrent ID of a transistor, and currents I1 and I2 in the switchingcircuit of the comparison example 2;

FIG. 26A is a graph for showing time elapse changes in a voltage VG of agate electrode, a voltage VD of a drain electrode D of a transistor, andvoltages V1 and V2 in a switching circuit of a comparison example 3; andFIG. 26B is a graph for indicating time elapse changes in a draincurrent ID of a transistor, and currents I1 and I2 in the switchingcircuit of the comparison example 3;

FIG. 27 is a circuit diagram for indicating a switching circuit whereonly a series circuit is provided;

FIG. 28 is a circuit diagram for showing a switching circuit accordingto a fifth embodiment of the present invention;

FIG. 29 is a circuit diagram for representing an equivalent circuit ofthe switching circuit of the fifth embodiment;

FIG. 30 is a graph for showing a temporal change of a current flowingthrough a diode when a transistor is turned OFF;

FIG. 31 is a graph for showing a temporal change of a gate voltage whenthe transistor is turned OFF;

FIG. 32 is a graph for showing a temporal change of a drain voltage whenthe transistor is turned OFF;

FIG. 33 is a graph for representing a relationship between a turn-OFFloss and a surge voltage;

FIG. 34 is the circuit diagram for showing the load driving circuitcontaining the surge voltage suppressing circuit;

FIG. 35 is the graph for representing the temporal change of the drainvoltage when the conventional load driving circuit is turned OFF;

FIG. 36A is a sectional view for showing a major portion of asemiconductor substrate where a transistor, a diode, and a capacitor ofthe switching circuit according to the fifth embodiment have beenformed; and FIG. 36B is a circuit diagram for indicating an equivalentcircuit thereof;

FIG. 37 is a sectional view for showing a major portion of thesemiconductor substrate where a transistor, a diode, and a capacitor asto an modification of the switching circuit according to the fifthembodiment have been formed;

FIG. 38 is a sectional view for showing a major portion of thesemiconductor substrate where a transistor, a diode, and a capacitor asto another modification of the switching circuit according to the fifthembodiment have been formed;

FIG. 39 is a sectional view for showing a major portion of thesemiconductor substrate where a transistor, a diode, and a capacitor asto another modification of the switching circuit according to the fifthembodiment have been formed;

FIG. 40 is a sectional view for showing a major portion of thesemiconductor substrate where a transistor, a diode, and a capacitor asto another modification of the switching circuit according to the fifthembodiment have been formed;

FIG. 41 is a sectional view for showing a major portion of thesemiconductor substrate where a transistor, a diode, and a capacitor asto another modification of the switching circuit according to the fifthembodiment have been formed;

FIG. 42A is a sectional view for showing a major portion of asemiconductor substrate where a transistor, a diode, and a capacitor asto one modification of the switching circuit according to the fifthembodiment have been formed; and FIG. 42B is a circuit diagram forindicating an equivalent circuit thereof;

FIG. 43 is a sectional view for showing a major portion of thesemiconductor substrate where a transistor, a diode, and a capacitor asto another modification of the switching circuit according to the fifthembodiment have been formed;

FIG. 44 is a sectional view for showing a major portion of thesemiconductor substrate where a transistor, a diode, and a capacitor asto another modification of the switching circuit according to the fifthembodiment have been formed;

FIG. 45 is a sectional view for showing a major portion of thesemiconductor substrate where a transistor, a diode, and a capacitor asto another modification of the switching circuit according to the fifthembodiment have been formed;

FIG. 46 is a circuit diagram for showing a basic arrangement of adriving circuit according to a sixth embodiment of the presentinvention;

FIG. 47A to FIG. 47E are graphs for indicating operation waveformdiagrams of the driving circuit of the sixth embodiment;

FIG. 48 is a circuit diagram for showing a concrete arrangement of thedriving circuit according to the sixth embodiment of the presentinvention;

FIG. 49A to FIG. 49C are sectional views for indicating conditions ofdepletion layers which are expanded/compressed in a pinch resistor ofthe sixth embodiment;

FIG. 50 is a graph for showing a relationship between a resistance valueof the pinch resistor and a voltage applied to the pinch resistor of thesixth embodiment;

FIG. 51 is a graph for showing a simulation result of the drivingcircuit of the sixth embodiment;

FIG. 52 is a graph for showing a simulation result of a driving circuitof a comparison example;

FIG. 53 is a graph for indicating a relationship between a surge voltageand a turn-OFF loss of a driving circuit according to a sixth embodimentof the present invention;

FIG. 54 is a circuit diagram for indicating a concrete arrangement ofone modification as to the driving circuit of the sixth embodiment;

FIG. 55 is a circuit diagram for indicating a concrete arrangement ofanother modification as to a driving circuit of a seventh embodiment ofthe present invention;

FIG. 56 is a graph for showing a relationship between a resistance valueof a pinch resistor and a voltage applied to the pinch resistor as to amodification of the seventh embodiment;

FIG. 57 is a graph for showing a simulation result of a driving circuitof a modification of the seventh embodiment;

FIG. 58 is a graph for indicating a relationship between a surge voltageand a turn-OFF loss of a driving circuit according to a modification ofthe seventh embodiment of the present invention;

FIG. 59 is a perspective view for indicating a pinch resistor of aneighth embodiment of the present invention;

FIG. 60 is a plan view for representing a surface layout of asemiconductor substrate according to the eighth embodiment;

FIG. 61 is a sectional view for showing the semiconductor substrate ofthe eighth embodiment, taken along a line LXI-LXI of FIG. 60;

FIG. 62 is a sectional view for showing the semiconductor substrate ofthe eighth embodiment, taken along a line LXII-LXII of FIG. 60;

FIG. 63 is a plan view for representing a n example of a zener diodeaccording to a ninth embodiment of the present invention;

FIG. 64 is a plan view for representing another example of the zenerdiode according to the ninth embodiment of the present invention;

FIG. 65A to FIG. 65E are the graphs for representing operation waveformdiagrams of the conventional driving circuit;

FIG. 66 is a circuit diagram for showing an arrangement of a drivingcircuit according to a tenth embodiment of the present invention;

FIG. 67A to FIG. 67E are graphs for showing operation waveform diagramsof a transistor employed in the tenth embodiment;

FIG. 68 is a circuit diagram for indicating an arrangement of a drivingcircuit with employment of a p-MOSFET;

FIG. 69 is a graph for indicating a variation of a drain-to-sourcevoltage of a transistor which is driven by the driving circuit using thep-MOSFET;

FIG. 70 is a graph for representing a relationship between a surgevoltage and a turn-OFF loss of a transistor which is driven by thedriving circuit using the p-MOSFET;

FIG. 71 is a graph for indicating a variation of a drain-to-sourcevoltage of a transistor when a resistance value of a second resistor ischanged in the driving circuit using the p-MOSFET;

FIG. 72 is a graph for representing a relationship between a surgevoltage and a turn-OFF loss of a transistor when the resistance value ofthe second resistor is changed in the driving circuit using thep-MOSFET;

FIG. 73 is a graph for indicating a variation of a drain-to-sourcevoltage of a transistor when a threshold value of the p-MOSFET ischanged in the driving circuit using the p-MOSFET;

FIG. 74 is a graph for representing a relationship between a surgevoltage and a turn-OFF loss of a transistor when the threshold value ofthe p-MOSFET is changed in the driving circuit using the p-MOSFET;

FIG. 75 is a circuit diagram for showing an arrangement of a drivingcircuit with employment of an n-MOSFET;

FIG. 76 is a graph for indicating a variation of a drain-to-sourcevoltage of a transistor when a resistance value of a second resistor ischanged in the driving circuit using the n-MOSFET;

FIG. 77 is a graph for representing a relationship between a surgevoltage and a turn-OFF loss of a transistor when the resistance value ofthe second resistor is changed in the driving circuit using then-MOSFET;

FIG. 78 is a graph for indicating a variation of a drain-to-sourcevoltage of a transistor when a threshold value of the n-MOSFET ischanged in the driving circuit using the n-MOSFET; and

FIG. 79 is a graph for representing a relationship between a surgevoltage and a turn-OFF loss of a transistor when the threshold value ofthe n-MOSFET is changed in the driving circuit using the n-MOSFET.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First Embodiment

FIG. 1 shows a major portion of a switching circuit 1 according to afirst embodiment of the present disclosure. The switching circuit 1 hasbeen equipped with a DC power supply PS, a load resistor R1, and atransistor (namely, MOSFET) 5, while these circuit elements has beenseries-connected to each other. The switching circuit 1 has also beequipped with a capacitance component C1, a resistance component R2, anda stray inductance component L which is caused by a wiring line, and thelike. When a current flowing between main electrodes (namely, drain Dand source S) of the transistor 5 is rapidly changed by the strayinductance component L, a large surge voltage is generated between themajor electrodes of the transistor 5, and thus, there are somepossibilities that this large surge voltage may damage the transistor 5,or may become noise which may give an adverse influence to otherappliances.

The transistor 5 is a unipolar type transistor, while a drain D thereofhas been connected to the high potential side and a source S thereof hasbeen connected to the low potential side. A gate voltage control circuit3 has been connected to a gate G of the transistor 5. The gate voltagecontrol circuit 3 outputs such a gate voltage. This gate voltage isinverted at a high frequency between a potential which causes thetransistor 5 to become conductive, and another potential which causesthe transistor 5 to become non-conductive.

A signal path between the drain D and the gate G of the transistor 5 hasbeen connected by a series circuit constructed of a zener diode 9 and acapacitor 7. A cathode of the zener diode 9 has been connected to thedrain D of the transistor 5, and an anode of the zener diode 9 has beenconnected via a capacitor 7 to the gate G of the transistor 5. While thetransistor 5 is turned OFF, a potential of the drain D is higher than apotential of the gate G, so that a reverse bias voltage is applied tothe zener diode 9.

FIG. 2 shows a temporal change of a drain voltage “Vd” generated beforeand after the transistor 5 of FIG. 1 is turned OFF. In this example, adrain voltage “Vd” implies a voltage between the drain D and the sourceS of the transistor 5 (namely, drain-to-source voltage). Since thesource voltage is defined as a reference voltage (zero volt) in thebelow-mentioned specification, a drain-to-source voltage will bereferred to as a drain voltage “Vd.” An abscissa of FIG. 2 indicates atime (μ second), and an ordinate thereof represents a voltage. In FIG.2, at a time instant of 1.53μ seconds, the transistor 5 has been turnedOFF. When the transistor 5 is turned ON, the drain voltage Vd isincreased toward a power supply voltage (in this case, 100 V).

FIG. 3 to FIG. 5 indicate temporal changes of drain voltages “Vd” anddrain currents “Id”, which are produced before and after the transistor5 of FIG. 1 is turned OFF. In this case, a drain current “id” implies acurrent flowing between the drain D and the source S of the transistor 5(drain-to-source current). Abscissas of FIG. 3 to FIG. 5 show a time (μsecond), left ordinates indicate currents, and right ordinates representvoltages. Even in FIG. 3 to FIG. 5, at a time instant of 1.53μ seconds,the transistor 5 has been turned OFF. When the transistor 5 is turnedOFF, the drain current Id is decreased toward zero.

A curve of “Vd-1” of FIG. 2 indicates a temporal change of a drainvoltage Vd of a comparison example in order to be compared with that ofthe first embodiment. The switching circuit of the comparison examplecorresponds to such a case that the series circuit constructed of thezener diode 9 and the capacitor 7 is not present between the drain D andthe gate G of the transistor 5. In the switching circuit of thecomparison example, the drain voltage Vd exceeds the power supplyvoltage (100 volt.) and is largely increased, and thereafter, isconverged to the power supply voltage while the drain voltage isoscillating. A surge voltage Vs may be defined by a difference between amaximum value of such a drain voltage Vd gene rated after the transistor5 is turned OFF and a drain voltage Vd which becomes stable after thetransistor 5 is turned OFF. In the switching circuit of the comparisonexample, as shown in the curve of Vd-1 of FIG. 2, a large surge voltage“Vs-1” is generated.

In FIG. 3, a temporal change of the drain current Id of the switchingcircuit of the comparison example has been described in combination withthe temporal change of the drain voltage.

A switching loss is equal to such a value obtained by integratingabsolute values of drain voltages “Vd” X drain currents “Id” as to atime period after the transistor 5 has been turned OFF until the draincurrent Id is stabilized at a stationary value.

In the switching circuit of the comparison example, since both the drainvoltage Vd and the drain current Id are rapidly changed, a switchingloss is small.

FIG. 6 represents a simulation result of a switching loss (abscissa) anda surge voltage (ordinate). A point “VIA” of FIG. 6 indicates asimulation result in the case where a signal path between a drain D anda gate G of a MOSFET is opened, and indicates such a fact that a surgevoltage is high, as compared with results 2 and 3 of this firstembodiment (will be explained later).

A hyperbolic curve shown in FIG. 6 represents a simulation result as toswitching losses and surge voltages in the case that various sorts ofMOSFETs are employed, while these MOSFETs are commercially available atthe present stage. A clear trade-off relationship between the switchinglosses and the surge voltages is present.

The point “VIA” is located on the trade-off curve.

A curve of “Vd-2” shown in FIG. 2 represents a temporal change of adrain voltage Vd in such a case that the signal path between the drain Dand the gate G of the transistor 5 is connected by a series circuitconstituted by a zener diode 9 having a zener voltage of 90 V and acapacitor 7 having a capacitance of 1 nF. As apparent from this curve, achanging speed of the drain voltage Vd is changed before and after apoint “P1.” Before this point “P1”, the drain voltage Vd is rapidlychanged, whereas the change of the drain voltage Vd is slowed after thepoint P1. In this case, the point “P1” corresponds to such a point thatthe drain-to-gate voltage becomes 90 V, so that the zener diode 9 breaksdown.

When the zener diode 9 breaks down at the point P1, a gate voltage(correctly speaking, although this voltage is gate-to-source voltage,since source voltage constitutes reference voltage, this voltage issimply referred to as “gate voltage”) receives the influence caused bythe drain voltage, so that this voltage is increased. The increasinggate voltage may alternatively delay the turn-OFF speed of thetransistor 5, and may alternatively slow the increasing speed of thedrain voltage Vd and the decreasing speed of the drain current Id so asto suppress the surge voltage to a lower surge voltage.

After the point P1, the change of the drain voltage Vd is slowed, sothat a surge voltage “Vs-2” (refer to FIG. 4) may be suppressed.

In FIG. 4, the below-mentioned temporal change is also described incombination with the temporal change of the drain voltage Vd, namely thetemporal change of the drain current “Id” in the case that the seriescircuit is employed which is constituted by the zener diode 9 having thezener voltage of 90 V, and the capacitor 7 having the capacitance of 1nF is also described.

In such a case that the signal path between the drain D and the gate Gof the transistor 5 is connected by the series circuit constituted bythe zener diode 9 having the zener voltage of 90 V and the capacitor 7having the capacitance of 1 nF, both a change of the drain voltage Vdand a change of the drain current Id are slowed after the point P1. As aresult, a switching loss is slightly increased. However, before thepoint P1, an influence of the capacitor 7 does not appear, but both thedrain voltage Vd and the drain current Id are rapidly changed. As aconsequence, an increased amount of the switching loss may be maintainedwithin a small amount range.

A point “VIB” of FIG. 6 shows a simulation result in such a case thatthe signal path between the drain D and the gate G of the transistor 5is connected by the series circuit constituted by the zener diode 9having the zener voltage of 90 V and the capacitor 7 having thecapacitance of 1 nF.

Although a switching loss of this point VIB is slightly increased, ascompared with that of the point VIA, a surge voltage is remarkablysuppressed. While comparing with the conventional circuit in which thesignal paths between the drains and the gates of the various sorts ofpresently available MOSFETs are opened, when the serge voltage Vs isequivalently maintained, the switching loss can be suppressed to asmaller switching loss than the conventional switching loss, whereaswhen the switching loss is equivalently maintained, the surge voltage Vscan be suppressed to a lower surge voltage than the conventional surgevoltage.

A curve of “Vd-3” shown in FIG. 2 represents a temporal change of adrain voltage Vd in such a case that the signal path between the drain Dand the gate G of the transistor 5 is connected by a series circuitconstituted by a zener diode 9 having a zener voltage of 50 V and acapacitor 7 having a capacitance of 1 nF. As apparent from this curve, achanging speed of the drain voltage Vd is changed before and after apoint “P2.” Before this point “P2”, the drain voltage Vd is rapidlychanged, whereas the change of the drain voltage Vd is slowed after thepoint P2. In this case, the point “P2” corresponds to such a point thatthe drain-to-gate voltage becomes 50 V, so that the zener diode 9 breaksdown.

After the point P2, the change of the drain voltage Vd is slowed, sothat a surge voltage “Vs-3” (refer to FIG. 5) may be suppressed.

In FIG. 5, the below-mentioned temporal change is also described incombination with the temporal change of the drain voltage, namely thetemporal change of the drain current “Id” in the case that the seriescurrent is employed which is constituted by the zener diode 9 having thezener voltage of 50 V, and the capacitor 7 having the capacitance of 1nF is also described.

In such a case that the signal path between the drain D and the gate Gof the transistor 5 is connected by the series circuit constituted bythe zener diode 9 having the zener voltage of 50 V and the capacitor 7having the capacitance of 1 nF, both a change of the drain voltage Vdand a change of the drain current Id are slowed after the point P2. As aresult, a switching loss is increased. However, before the point P2, aninfluence of the capacitor 7 does not appear, but both the drain voltageVd and the drain current Id are rapidly changed. As a consequence, anincreased amount of the switching loss is maintained within a smallamount range.

A point “VIC” of FIG. 6 shows a simulation result in such a case thatthe signal path between the drain D and the gate G of the transistor 5is connected by the series circuit constituted by the zener diode 9having the zener voltage of 50 V and the capacitor 7 having thecapacitance of 1 nF.

As compared with the point “VIB”, although a switching loss of thispoint “VIC” is slightly increased, a surge voltage is furthermoresuppressed.

In the case that such a zener is employed whose zener voltage isapproximately 0.5 to 1.0 times higher than a voltage applied between adrain and a source of a transistor when the transistor is turned OFF(implies that transistor is stabilized under OFF state), the followingfact can be revealed. That is, both a surge voltage and a switching losscan be suppressed under better balance condition. In the case that thesurge voltage is strongly suppressed, it is preferable to utilize azener diode having a lower zener voltage. In the case that an increaseof the switching loss is strongly suppressed, it is desirable to employa zener diode having a higher zener voltage.

A curve of “Vd-4” shown in FIG. 2 represents a temporal change of adrain voltage Vd in such a case that the signal path between the drain Dand the gate G of the transistor 5 is connected by a capacitor 7 havinga capacitance of 1 nF. In this case, the zener diode 9 is not used. Asapparent from the curve, during a time period from a commencement ofturning-OFF of the transistor 5 until the transistor 5 is stabilizedunder OFF state, the change of the drain voltage Vd is slow. As aresult, although the surge voltage Vs is suppressed, the switching lossis largely increased.

A point “VID” of FIG. 6 shows a simulation result in such a case thatthe signal path between the drain D and the gate G of the transistor 5is connected by the capacitor 7 having the capacitance of 1 nF.

A switching loss of this point VID is increased, as compared with theswitching losses of the points VIB and VIC. It can be understood thatthe utilization of the zener diode 9 becomes effective.

It should also be noted that it is preferable that the stray capacitanceof the zener diode 9 is small. If the stray capacitance of the zenerdiode 9 is small, then both a voltage change and a current change justafter the transistor 5 is turned OFF can be kept at high speeds. If sucha zener diode 9 having a small stray capacitance is employed, then it ispossible to suppress a switching loss to a small value.

Second Embodiment

FIG. 7 shows a major portion of a switching circuit 10 according to asecond embodiment of the present disclosure. The switching circuit 10has been equipped with a DC power supply PS, a load resistor R1, and atransistor (namely, MOSFET) 5, while these circuit elements has beenseries-connected to each other. The switching circuit 10 has also beequipped with a capacitance component C1, a resistance component R2, anda stray inductance component L which is caused by a wiring line, and thelike. When a current flowing between main electrodes (namely, drain Dand source S) of the transistor 5 is rapidly changed by the strayinductance component L, a large surge voltage is generated between themajor electrodes of the transistor 5, and thus, there are somepossibilities that this large surge voltage may damage the transistor 5,or may become noise which may give an adverse influence to otherappliances.

The transistor 5 is a unipolar type transistor, while a drain D thereofhas been connected to the high potential side and a source S thereof hasbeen connected to the low potential side. A gate voltage control circuit3 has been connected to a gate G of the transistor 5. The gate voltagecontrol circuit 3 outputs such a gate voltage. This gate voltage isinverted at a high frequency between a potential which causes thetransistor 5 to become conductive, and another potential which causesthe transistor 5 to become non-conductive.

A signal path between the drain D and the source S of the transistor 5has been connected by a series circuit constructed of a zener diode 9and a capacitor 7. A cathode of the zener diode 9 has been connected tothe drain D of the transistor 5, and an anode of the zener diode 9 hasbeen connected via a capacitor 7 to the source S of the transistor 5.While the transistor 5 is turned OFF, a potential of the drain D ishigher than a potential of the source S, so that a reverse bias voltageis applied to the zener diode 9.

FIG. 8 shows a temporal change of a drain voltage “Vd” generated beforeand after the transistor 5 of FIG. 7 is turned OFF. In this example, adrain voltage “Vd” implies a voltage between the drain D and the sourceS of the transistor 5 (namely, drain-to-source voltage). An abscissa ofFIG. 8 indicates a time (μ second), and an ordinate thereof represents avoltage. In FIG. 8, at a time instant of 1.53μ seconds, the transistor 5has been turned OFF. When the transistor 5 is turned ON, the drainvoltage Vd is increased toward a power supply voltage (in this case, 100V).

FIG. 9 to FIG. 10 indicate temporal changes of drain voltages “Vd” anddrain currents “Id”, which are produced before and after the transistor5 of FIG. 7 is turned OFF. In this case, a drain current “id” implies acurrent flowing between the drain D and the source S of the transistor 5(drain-to-source current). Abscissas of FIG. 9 to FIG. 10 show a time (μsecond), left ordinates indicate currents, and right ordinates representvoltages. Even in FIG. 9 to FIG. 10, at a time instant of 1.53μ seconds,the transistor 5 has been turned OFF. When the transistor 5 is turnedOFF, the drain current Id is decreased toward zero.

A curve of “Vd-1” of FIG. 8 indicates a temporal change of a drainvoltage Vd of a comparison example in order to be compared with that ofthe second embodiment. The switching circuit of the comparison examplecorresponds to such a case that the series circuit constructed of thezener diode 9 and the capacitor 7 is not present between the drain D andthe source S of the transistor 5. In the switching circuit of thecomparison example, the drain voltage Vd exceeds the power supplyvoltage (100 volt.) and is largely increased, and thereafter, isconverged to the power supply voltage while the drain voltage isoscillating. A surge voltage Vs may be defined by a difference between amaximum value of such a drain voltage Vd generated after the transistor5 is turned OFF and a drain voltage Vd which becomes stable after thetransistor 5 is turned OFF. In the switching circuit of the comparisonexample, as shown in the curve of Vd-1 of FIG. 8, a large surge voltage“Vs-1” is generated.

In the switching circuit of the comparison example, since both the drainvoltage Vd and the drain current Id are rapidly changed, a switchingloss is small.

FIG. 11 represents a simulation result of a switching loss (abscissa)and a surge voltage (ordinate). A point “XIA” of FIG. 11 indicates asimulation result in the case where a signal path between a drain D anda source S of a MOSFET is opened, and indicates such a fact that a surgevoltage is high, as compared with results 5 to 7 of this secondembodiment (will be explained later).

A hyperbolic curve shown in FIG. 11 corresponds to a trade-off curveshown in FIG. 6.

A curve of “Vd-5” shown in FIG. 8 represents a temporal change of adrain voltage Vd in such a case that the signal path between the drain Dand the source S of the transistor 5 is connected by a series circuitconstituted by a zener diode 9 having a zener voltage of 90 V and acapacitor 7 having a capacitance of 1 nF. As apparent from this curve, achanging speed of the drain voltage Vd is changed at a stage when thedrain voltage Vd is increased to the power supply voltage. The drainvoltage Vd is rapidly changed just after the transistor 5 is turned OFF,whereas the changing speed of the drain voltage Vd is slowed after sucha timing when the drain voltage Vd is approximated to the power supplyvoltage. At timing when the changing speed of the drain voltage Vd ischanged corresponds to such a time instant that the drain-to-sourcevoltage becomes 90 V, so that the zener diode 9 breaks down. When thezener diode 9 breaks down, the changing speed of the drain voltage Vd isslowed. As a result, the surge voltage Vs is suppressed.

A point “XIB” of FIG. 11 shows a simulation result in such a case thatthe signal path between the drain D and the source S of the transistor 5is connected by the series circuit constituted by the zener diode 9having the zener voltage of 90 V and the capacitor 7 having thecapacitance of 1 nF.

Although a switching loss of this point VIB is slightly increased, ascompared with that of the point XIA, a surge voltage is suppressed.While comparing with the conventional circuit in which the signal pathsbetween the drains and the sources of the various sorts of presentlyavailable MOSFETs are opened, when the surge voltage Vs is equivalentlymaintained, the switching loss can be suppressed to a smaller switchingloss than the conventional switching loss, whereas when the switchingloss is equivalently maintained, the surge voltage Vs can be suppressedto a lower surge voltage than the conventional surge voltage. However,in such a case that the capacitor 7 is arranged between the source S andthe drain D, the following fact can be revealed: That is, if thecapacitance of this capacitor 7 is approximately 1 nF, then thesuppression effect of the surge voltage cannot become conspicuous.

A curve of “Vd-6” shown in FIG. 8 represents a temporal change of adrain voltage Vd in such a case that a signal path between the drain Dand the source S of the transistor 5 is connected by a series circuitconstituted by a zener diode 9 having a zener voltage of 90 V and acapacitor 7 having a capacitance of 3 nF. As apparent from this curve, achanging speed of the drain voltage Vd is changed at a stage when thedrain voltage Vd is increased to the power supply voltage. The drainvoltage Vd is rapidly changed just after the transistor 5 is turned OFF,whereas the changing speed of the drain voltage Vd is slowed after sucha timing when the drain voltage Vd is approximated to the power supplyvoltage. At timing when the changing speed of the drain voltage Vd ischanged corresponds to such a time instant that the drain-to-sourcevoltage becomes 90 V, so that the zener diode 9 breaks down. When thezener diode 9 breaks down, the changing speed of the drain voltage Vd isslowed. As a result, a surge voltage “Vs-6” (refer to FIG. 9) issuppressed. If a capacitance of a capacitor connected between the drainD and the source S is increased from 1 nF to 3 nF, then the changingspeed of the drain voltage Vd after the time instant when the zenerdiode 9 breaks down is furthermore slowed. The surge voltage “V3-6”(refer to FIG. 9) may be furthermore suppressed to a low surge voltage.

In FIG. 9, the below-mentioned temporal change is also described incombination with the temporal change of the drain voltage, namely thetemporal change of the drain current “Id” in the case that the seriescurrent is employed which is constituted by the zener diode 9 having thezener voltage of 90 V, and the capacitor 7 having the capacitance of 3nF is also described.

In such a case that the signal path between the drain D and the source Sof the transistor 5 is connected by the series circuit constituted bythe zener diode 9 having the zener voltage of 90 V and the capacitor 7having the capacitance of 3 nF, both a change of the drain voltage Vdand a change of the drain current Id are slowed after a point P3 wherethe zener diode 9 breaks down. As a result, a switching loss isincreased. However, before the point P3, an influence of the capacitor 7does not appear, but both the drain voltage Vd and the drain current Idare rapidly changed. As a consequence, an increased amount of theswitching loss is maintained within a small amount range.

A point “XIC” of FIG. 11 shows a simulation result in such a case thatthe signal path between the drain D and the source S of the transistor 5is connected by the series circuit constituted by the zener diode 9having the zener voltage of 90 V and the capacitor 7 having thecapacitance of 3 nF.

As compared with the point “XIA”, although a switching loss of thispoint “XIC” is slightly increased, a surge voltage is conspicuouslysuppressed.

A curve of “Vd-7” shown in FIG. 8 represents a temporal change of adrain voltage Vd in such a case that the signal path between the drain Dand the gate G of the transistor 5 is connected by a series circuitconstituted by a zener diode 9 having a zener voltage of 50 V and acapacitor 7 having a capacitance of 3 nF. As apparent from this curve, achanging speed of the drain voltage Vd is changed in a half way when thedrain voltage Vd is changed. The drain voltage Vd is rapidly changedjust after the transistor 5 is turned OFF, whereas the changing speed ofthe drain voltage Vd is slowed in a half way when the drain voltage Vdis increased toward the power supply voltage. At timing when thechanging speed of the drain voltage Vd is changed corresponds to such atime instant that the drain-to-source voltage becomes 50V, so that thezener diode 9 breaks down. When the zener diode 9 breaks down, thechanging speed of the drain voltage Vd is slowed. As a result, a surgevoltage “Vs-7” (refer to FIG. 10) is suppressed to a low surge voltage.

In FIG. 10, the below-mentioned temporal change is also described incombination with the temporal change of the drain voltage, namely thetemporal change of the drain current “Id” in the case that the seriescurrent is employed which is constituted by the zener diode 9 having thezener voltage of 50 V, and the capacitor 7 having the capacitance of 3nF is also described.

In such a case that the signal path between the drain D and the source Sof the transistor 5 is connected by the series circuit constituted bythe zener diode 9 having the zener voltage of 50 V and the capacitor 7having the capacitance of 3 nF, both a change of the drain voltage Vdand a change of the drain current Id are slowed after a point P4 whenthe zener diode 9 breaks down. As a result, a switching loss isincreased. However, before the point P4, an influence of the capacitor 7does not appear, but both the drain voltage Vd and the drain current Idare rapidly changed. As a consequence, an increased amount of theswitching loss is maintained within a small amount range.

A point “XID” of FIG. 11 shows a simulation result in such a case thatthe signal path between the drain D and the source S of the transistor 5is connected by the series circuit constituted by the zener diode 9having the zener voltage of 50 V and the capacitor 7 having thecapacitance of 3 nF.

As compared with the point “XIC”, although a switching loss of thispoint “XID” is slightly increased, a surge voltage “Vs-7” (refer to FIG.10) is furthermore suppressed.

Also, when the series circuit is employed between the source S and thedrain D of the transistor 5, in the case that such a zener is employedwhose zener voltage is approximately 0.5 to 1.0 times higher than avoltage applied between a drain and a source of a transistor when thetransistor is turned OFF (implies that transistor is stabilized underOFF state), the following fact can be revealed. That is, both a surgevoltage and a switching loss can be suppressed under better balancecondition. In the case that the surge voltage is strongly suppressed, itis preferable to utilize a zener diode having a lower zener voltage. Inthe case that an increase of the switching loss is strongly suppressed,it is desirable to employ a zener diode having a higher zener voltage.Also, when the surge voltage is strongly suppressed, it is preferable toutilize a capacitor having a large capacitance. When the switching lossis strongly suppressed, it is preferable to utilize a capacitor having asmall capacitance.

A curve of “Vd-8” shown in FIG. 8 represents a temporal change of adrain voltage Vd in such a case that the signal path between the drain Dand the source S of the transistor 5 is connected by a capacitor 7having a capacitance of 3 nF. In this case, the zener diode 9 is notused. As apparent from the curve, during a time period from acommencement of turning-OFF of the transistor 5 until the transistor 5is stabilized under OFF state, the change of the drain voltage Vdbecomes slow. As a result, although the surge voltage Vs is suppressed,the switching loss is largely increased.

A point “XIE” of FIG. 11 shows a simulation result in such a case thatthe signal path between the drain D and the source S of the transistor 5is connected by the capacitor 7 having the capacitance of 3 nF.

A switching loss of this point XIE is increased, as compared with theswitching losses of the points XIB to XID. It can be understood that theutilization of the zener diode 9 becomes effective.

It should also be noted that it is preferable that even when thecapacitor 7 is employed between the drain D and the source S of thetransistor 5, the stray capacitance of the zener diode 9 is small. Ifthe stray capacitance of the zener diode 9 is small, then both a voltagechange and a current change just after the transistor 5 is turned OFFcan be kept at high speeds. If such a zener diode 9 having a small straycapacitance is employed, then it is possible to suppress a switchingloss to a small value.

Also, in any of the first embodiment and the second embodiment, thearranging order of the zener diode 9 and the capacitor 7 is changed,equivalent circuits may be obtained and the same effect may be achieved.

Alternatively, the first embodiment and the second embodiment may becarried out at the same time.

Also, the above-described embodiments are not limited only to a unipolartransistor such as a MOSFET, but may alternatively employ a bipolartransistor such as an IGBT by which a similar effect may be achieved.Since the circuit of the present disclosure is additionally provided,both the surge voltage and the switching loss may be suppressed at thesame time.

A transistor, and a series circuit made of a zener diode and acapacitor, which connects a signal path between a drain (or collector)and a gate of the transistor, or another signal path between the drain(or collector) and a source (or emitter) thereof, may be alternativelyformed in an integral body on the same semiconductor substrate. Aconcrete semiconductor mode will be described in a third embodiment anda fourth embodiment. In this case, an extra component is no longerrequired, and both a surge voltage and a switching loss may besuppressed by one chip.

In accordance with the present disclosure, even when the conventionaltransistor is directly employed, since such low cost general-purposecomponents as a zener diode and a capacitor are utilized, both a surgevoltage and a switching loss may be suppressed. Also, a capacitance of acapacitor and a zener voltage of a zener diode are changed, so thatvarious sorts of tuning can be realized. For instance, it is possible torealize such a characteristic capable of suppressing a switching loss tothe minimum switching loss within an allowable surge voltage.

Third Embodiment

FIG. 12A to FIG. 16B indicate examples in which transistors, zenerdiodes 9, and capacitors 7, which have been employed in switchingcircuits, have been formed on the same semiconductor substrate 20. Anyof the examples shown in FIG. 12A to FIG. 16B are to embody the portionsof the transistor, the zener diode 9, and the capacitor 7 employed inthe switching circuit 1 of the first embodiment. Any of these examplesshown in FIG. 12A to FIG. 16B corresponds to such a semiconductor modethat while the transistor and the capacitor 7 have been formed in thesemiconductor substrate 20, the zener diode 9 has been provided on thesemiconductor substrate 20. The transistor, the zener diode 9, and thecapacitor 7 have been formed in an integral body by utilizing thesemiconductor substrate 20, and have been constituted in a single chip.It should also be understood that single crystal silicon has beenemployed in the semiconductor substrate 20.

The semiconductor mode shown in FIG. 12A and FIG. 12B indicate such anexample that the series circuit constructed of the zener diode 9 and thecapacitor 7 has been connected between a drain D and a gate G of thetransistor, and also, the capacitor 7 has been inserted between thedrain D of the transistor and the zener diode 9. FIG. 12A schematicallyindicates a sectional view of a major portion of the semiconductorsubstrate 20. FIG. 12B shows an equivalent circuit of the structureshown in FIG. 12A.

FIG. 12A indicates a vicinity of a boundary between an element regionand a periphery region. The element region corresponds to such a regionthat a semiconductor region group has been formed which is required inorder to temporally switch a signal path between the major electrodes(namely, drain D and source S) under conductive state and undernon-conductive state. The element region has been arranged at a centerside of the semiconductor substrate 20. The periphery region has beenarranged around a central region in a circular manner. A terminal regionhas been formed in the periphery region. The terminal region has beenarranged around the central region in the circular manner. The terminalregion corresponds to such a region that when the transistor is underOFF state, a voltage applied to the transistor is loaded along a lateraldirection by expanding a depletion layer from the element region towarda side direction. Although a detailed description will be made later, adimension ratio of widths of the terminal region along the lateraldirection indicated in FIG. 12A is represented in a compression mode forthe sake of clear illustrations.

The semiconductor substrate 20 has been provided with an n⁺ type drainregion 22 and an n⁻ type drift region 24. The n⁺ type drain region 22has been formed on a rear surface of the semiconductor substrate 20,whereas the n⁻ type drift region 24 has been provided on the drainregion 22. Both the drain region 22 and the drift region 24 have beencontinuously provided along the lateral direction from the elementregion to the periphery region within the semiconductor substrate 20.

A p type body region 76, an n⁺ type source region 72, and a p⁺ type bodycontact region 74 have been provided on a front surface of thesemiconductor substrate 20 of the element region. A body region 76 hasbeen continuously provided on the front surface of the semiconductorsubstrate 20 of the element region along the lateral direction. Thesource region 72 and the body contact region 74 have been selectivelyprovided on the front surface of the semiconductor substrate 20. Thesource region 72 has been isolated from the drift region 24 by the bodyregion 76. The body contact region 74 contains an impurity in higherconcentration than that of the body region 76. Both the source region 72and the body contact region 74 have been electrically connected to thesource S provided on the semiconductor substrate 20.

A trench gate electrode 64 and a gate insulating film 62 have beenfurther formed on the front surface of the semiconductor substrate 20 ofthe element region. The trench electrode 64 has been extended from thefront surface of the semiconductor substrate 20 toward the rear surfacethereof, and has penetrated through the body region 76 and then hasreached the drift region 24. The gate insulating film 62 has covered thetrench gate electrode 64 so as to electrically insulate the trench gateelectrode 64 from the semiconductor substrate 20. The trench gateelectrode 64 is located opposite to the body region 76 via the gateinsulating film 62, while the body region 76 has isolated the sourceregion 72 from the drift region 24. Poly crystalline silicon has beenemployed in the trench gate electrode 64. A silicon oxide has beenemployed in the gate insulating film 62.

A selective oxide film 42 (one example of insulating film) has beenprovided on the front surface of the semiconductor substrate 20 of theperiphery region. A silicon oxide has been employed in the selectiveoxide film 42. The selective oxide film 42 can relax an electric fieldof the front surface of the semiconductor substrate 20 of the terminalregion. Generally speaking, there are many possibilities that suchstructures as, for example, a RESURF layer and a guard ring have beenprovided within the semiconductor substrate 20 located under theselective oxide film 42. These structures have been provided in order torelax electric field concentration occurred in the peripheral portion ofthe element region, and thus, so as to improve a withstanding voltage ofthe transistor. In this third embodiment, for the sake of clearillustration, these structures have been omitted.

An n type diffusion semiconductor region 38 and an n⁺ type surfacediffusion semiconductor region 32 have been provided on a front surfaceof the semiconductor substrate 20 which is located in a further sidedirection than the terminal region. Both the n type diffusionsemiconductor region 38 and the n⁺ type surface diffusion semiconductor32 have been provided on at least a portion of the front surface of thesemiconductor substrate 20 located in the further side direction thanthe terminal region. In other words, the n type diffusion semiconductorregion 38 and the n⁺ type surface diffusion semiconductor region 32 neednot be provided along the periphery portion of the terminal region in acircular manner.

An embedded conductive region 36 and a coating insulation region 34 havebeen furthermore provided in a portion of the front surface of thesemiconductor substrate 20 which is located in a further side directionthan the terminal region. The embedded conductive region 36 has beenextended from the front surface of the semiconductor substrate 20 to therear surface thereof, and then has been stayed within the n typediffusion semiconductor region 38. The coating insulation region 34 hascovered the embedded conductive region 36 so as to electrically insulatethe embedded conductive region 36 from the semiconductor substrate 20.Poly crystalline silicon has been employed in the embedded conductiveregion 36. A silicon oxide has been employed in the coating insulationregion 34.

The embedded conductive region 36 has been located opposite to the ntype diffusion semiconductor region 38 via the coating insulation region34. As a consequence, the capacitor 7 has been constituted by theembedded conductive region 36, the coating insulation region 34, and then type diffusion semiconductor region 38. It should also be noted thatthe n type diffusion semiconductor region 38 has been formed in order tosuppress that a depletion layer is produced in a boundary between thecoating insulation region 34 and the n type diffusion semiconductorregion 38, and thus, so as to stabilize the capacitance of the capacitor7.

The embedded conductive region 36 has the same depth as that of thetrench gate electrode 64 of the element region. The coating insulationregion 34 has the same thickness as that of the gate insulating film 62of the element region. As a consequence, both the embedded conductiveregion 36 and the coating insulation region 34 can be formed at the sametime by utilizing the process for manufacturing the trench gateelectrode 64 of the element region and the gate insulating film 62.

The zener diode 9 has been formed on the front surface of the selectiveregion 42. The zener diode 9 has been provided with an anodesemiconductor region 54 containing a p type impurity, and a cathodesemiconductor region 52 containing an n type impurity. While the anodesemiconductor region 54 and the cathode semiconductor region 52 havebeen provided on the front surface of the selective oxide film 42, theanode semiconductor region 54 and the cathode semiconductor region 52are directly contacted thereto. As materials of the anode semiconductorregion 54 and the cathode semiconductor region 52, poly crystallinesilicon has been employed.

As represented in FIG. 12A, the anode semiconductor region 54 of thezener diode 9 has been electrically connected to the trench gateelectrode 64 of the transistor. The cathode semiconductor region 52 ofthe zener diode 9 has been electrically connected to the embeddedconductive region 36 of the capacitor 7. In other words, the cathodesemiconductor region 52 of the zener diode 9 has been electricallyconnected to one end of the capacitor 7. The n type diffusionsemiconductor region 38 has been electrically connected via the n⁺ typesurface diffusion semiconductor region 32 to the drain D. Also, the ntype diffusion semiconductor region 38 has been electrically connectedvia the drift region 24 and the drain region 22 to the drain D. As aconsequence, the other end of the capacitor 7 has been electricallyconnected to the drain D of the transistor. A switching circuit-purposechip for embodying an equivalent circuit shown in FIG. 12B has beenarranged by electrically connecting these electronic elements. It shouldalso be understood that these electric connections may be constitutedby, for example, aluminum wiring lines.

As previously described, if the zener diode 9 and the capacitor 7 areconstructed in the integral body on the same semiconductor substrate 20as the transistor, then the zener diode 9 and the capacitor 7 need notbe separately prepared. As a consequence, the switching circuit-purposechip disclosed in the specification of can be constructed withoutincreasing a total number of these components. If the transistor, thezener diode 9, and the capacitor 7 have been provided on the samesemiconductor substrate 20, then the switching circuit can be arrangedby employing a small number of the components. Thus, the switchingcircuit can be furthermore made compact, and can be manufactured withsuperior practical characteristics.

FIG. 13 and FIG. 14 indicate a technical idea for changing a capacitanceof the capacitor 7. The capacitance of the capacitor 7 is adjusted basedupon a thickness and an area of the coating insulation region 34. Thecapacitance of the capacitor 7 is inverse proportion to the thickness ofthe coating insulation region 34, and is direct proportion to the areaof the coating insulation region 34.

As indicated in FIG. 13, if the embedded conductive region 36 and thecoating insulation region 34 are formed up to a deep position of thesemiconductor substrate 20 so as to increase the area of the coatinginsulation region 34, then the capacitance of the capacitor 7 isincreased. On the other hand, as shown in FIG. 14, if the thickness ofthe coating insulation region 34 is increased, then the capacitance ofthe capacitor 7 is decreased. As previously explained, the semiconductormodes as to the embedded conductive region 36 and the coating insulationregion 34 are contrived, so that the capacitance of the capacitor 7 canbe readily adjusted.

FIG. 15 indicates such a semiconductor structural example that a superjunction structure has been formed in the drift region 24 of the elementregion. The drift region 24 of the super junction structure has beenformed by a repetition structure made of an n type column 24 acontaining an n type impurity, and a p type column 24 b containing a ptype impurity. The impurity concentration of the p type column 24 b issubstantially equal to that of the body region 76. In the super junctionstructure, when the transistor is turned OFF, a depletion layer isextended from the pn junction between the n type column 24 a and the ptype column 24 b, and thus, the drift region 24 can be substantiallycompletely depleted. As a result, if a pitch width between the n typecolumn 24 a and the p type column 24 b is adjusted, even when theimpurity concentration is increased, then the drift region 24 can besubstantially perfectly depleted. As a consequence, since the impurityconcentration of the n type column 24 a and the p type column 24 b canbe increased, the super junction structure is useful as such a techniquecapable of solving the trade-off relationship existed between theON-resistance (otherwise, ON voltage) of the transistor and thewithstanding voltage.

Generally speaking, a super junction structure is formed by utilizingthe semiconductor substrate 20 which contains an n type impurity inrelatively high concentration. Concretely speaking, a plurality oftrenches are formed from the front surface of the semiconductorsubstrate 20 by performing an anisotropic etching process, and then, ptype columns 24 b are crystal-grown within these trenches. The remainingportions when the plural trenches are formed become n type columns 24 a.As a result, such a super junction structure that the n type columns 24a and the p type columns 24 b are repeatedly arranged may bemanufactured.

In the case where the super junction structure is employed in the driftregion 24 of the element region, the semiconductor substrate 20 isemployed which contains the n type impurity in relatively highconcentration. As a consequence, in the case where the super junctionstructure is employed in the drift region 24 of the element region, theimpurity concentration of the drift region 24 of the periphery region isincreased. Accordingly, in this example, even if the n type diffusionsemiconductor region 38 is not provided which was formed in the case ofFIG. 12, then it is possible to suppress such a phenomenon that thedepletion layer is produced at the boundary of the coating insulationregion 34.

As previously described, the technique for forming the transistor, thezener diode 9, and the capacitor 7 on the same semiconductor substrate20 may be applied to various sorts of transistors. Generally speaking,there are many possibilities that a periphery region of a transistor hasbeen provided with the selective oxide film 42. If the zener diode 9 isprovided on the front surface of this selective oxide film 42, then thezener diode 9 may be provided above the semiconductor substrate 20 ofthe periphery region without any limitation to the sort of transistor.Since the space of the surface of the selective oxide film 42 isutilized, the zener diode 9 can be easily formed in the semiconductorsubstrate 20 in an integral body while increasing of the element area issuppressed. Also, if both the embedded conductive region 36 and thecoating insulation region 34 are utilized, then the capacitor 7 may beeasily formed within the semiconductor region 20 of the peripheryregion. Since both the embedded conductive region 36 and the coatinginsulation region 34 have the trench shapes, the capacitor 7 can beeasily formed in the semiconductor substrate 20 in an integral bodywhile increasing of the element area is suppressed.

Semiconductor modes shown in FIG. 16A and FIG. 16B are such anexemplification that the series circuit constructed of the zener diode 9and the capacitor 7 has been connected between the drain D and the gateG of the transistor, and further, the capacitor 7 has been insertedbetween the gate G of the transistor and the zener diode 9. FIG. 16A isa sectional view for schematically showing a major portion of thesemiconductor substrate 20. FIG. 16B indicates an equivalent circuithaving the semiconductor substrate of FIG. 16A.

As represented in FIG. 16A, in this semiconductor mode, a position ofthe capacitor 7 and a position of the zener diode 9 are located in anopposite arranging sense with those of FIG. 12A to FIG. 15. Also, in thesemiconductor mode of FIG. 16A, a p type diffusion semiconductor region138 and a p⁺ type contact region 132 have been provided on the frontsurface of the semiconductor substrate 20 of the periphery region. The ptype diffusion semiconductor region 138 surrounds both the embeddedconductive region 36 and the coating insulation region 34. The embeddedconductive region 36 and the coating insulation region 34 have beenisolated from the drift region 24 by the p type diffusion semiconductorregion 138. The embedded conductive region 36 and the p type diffusionsemiconductor region 138 have been located opposite to each other viathe coating insulation region 34. As a consequence, the p type diffusionsemiconductor region 138 surrounds both the embedded conductive region36 and the coating insulation region 34. The capacitor 7 has beenconstructed of the embedded conductive region 36, the coating insulationregion 34, and the p type diffusion semiconductor region 138.

As shown in FIG. 16A, the embedded semiconductor region 36 of thecapacitor 7 has been electrically connected to the gate G of thetransistor. In other words, one end of the capacitor 7 has beenelectrically connected to the gate G of the transistor. The p typediffusion semiconductor region 138 has been electrically connected viathe contact region 132 to the anode semiconductor region 54 of the zenerdiode 9. In other words, the other end of the capacitor 7 has beenelectrically connected to the anode semiconductor region 54 of the zenerdiode 9. The cathode semiconductor region 52 of the zener diode 9 hasbeen electrically connected to the drain D of the transistor. Aswitching circuit-purpose chip for embodying an equivalent circuit shownin FIG. 16B has been arranged by electrically connecting theseelectronic elements. It should also be understood that these electricconnections may be constituted by, for example, aluminum wiring lines.

As previously described, if the zener diode 9 and the capacitor 7 areconstructed in the integral body on the same semiconductor substrate 20as the transistor, then the zener diode 9 and the capacitor 7 need notbe separately prepared. As a consequence, the switching circuit-purposechip disclosed in the specification of the present disclosure can beconstructed without increasing a total number of these components. Ifthe transistor, the zener diode 9, and the capacitor 7 have beenprovided on the same semiconductor substrate 20, then the switchingcircuit can be arranged by employing a small number of the components.Thus, the switching circuit can be furthermore made compact, and can bemanufactured with superior practical characteristics.

Fourth Embodiment

FIG. 27 shows an example of a switching circuit 200 according to afourth embodiment of the present disclosure. The switching circuit 200has been equipped with a transistor 5, a control circuit 3, and a seriescircuit 6. The transistor 5 is employed by connecting a power supply“PS” and a load R1 in a series connecting manner between a drainelectrode “D” and a source electrode “S” of the transistor 5. Thecontrol circuit 3 is connected to a gate electrode “G” of the transistor5. The series circuit 6 is connected between the gate electrode G andthe drain electrode D of the transistor 5, and a capacitor 7 isseries-connected to a diode 9 a. It should be understood that referencenumeral “L” indicates a stray inductance. The switching circuit 200 cansuppress a surge voltage based upon the below-mentioned operation. Itshould also be noted that since a phenomenon which should be originallydescribed based upon formulae is explained by using an expression, thisexpression is not always perfect.

While the control circuit 3 are outputting a voltage for turning OFF thetransistor 5, a voltage (Vps−V_(F)) obtained by subtracting a forwarddirection voltage “V_(F)” of the diode 9 a from a power supply voltage“Vps” has being applied to the capacitor 7, and a charged voltage of thecapacitor 7 has become (Vps−V_(F)).

Next, when the control circuit 3 outputs a voltage for turning ON thetransistor 5, a voltage at the drain electrode D of the transistor 5 islowered. Since a reverse bias voltage is applied to the diode 9 a, thecapacitor 7 cannot be discharged, but maintains the charged voltage(Vps−V_(F)) which has been so far charged.

Next, when the control circuit 3 outputs a voltage for turning OFF thetransistor 5, the voltage at the drain electrode D of the transistor 5is increased. Since the stray inductance “L” is present, the voltage ofthe drain electrode D exceeds the power supply voltage Vps and isfurther increased. A so-called “surge voltage” is generated.

If the voltage at the drain electrode D of the transistor 5 exceeds asummed voltage between the charged voltage (Vps−V_(F)) of the capacitor7 and the forward direction voltage V_(F) of the diode 9 a, namelyexceeds the power supply voltage Vps, then the forward direction voltageis effected to the diode 9 a, and thus, a charging current starts toflow through the capacitor 7. As a result, since a discharging speed ofthe gate electrode G of the transistor 5 is slowed, a changing speed ofa drain current of the transistor 5 is slowed; and a changing speed of adrain current of the transistor 5 is slowed, so that a steep increase ofthe voltage at the drain electrode D of the transistor 5 can besuppressed, and thus, the surge voltage can be suppressed to a low surgevoltage.

It should also be noted that in the above-example, such a series circuithas been exemplified in which the capacitor 7 is arranged on the side ofthe gate electrode (control electrode), and the diode 9 a is arranged onthe side of the drain electrode (main electrode on the high voltageside). Alternatively, even if another series circuit in which the diode9 a is arranged on the gate electrode side and the capacitor 7 isarranged on the drain electrode side is employed, the surge voltage maybe suppressed to a lower voltage.

A breakdown voltage of a zener diode is fluctuated by approximately ±10%due to manufacturing tolerance, whereas when a voltage at a drainelectrode of a transistor exceeds the power supply voltage Vps, alowering speed of a voltage at a gate electrode of the transistor isslowed in accordance with a technique for utilizing a capacitor, and alowering speed of a drain current of the transistor is slowed. As aresult, a switching circuit which has a fluctuation as to a suppressioncapability of a surge voltage can be mass-produced.

As previously described, if the capacitor is utilized, then when thevoltage at the drain electrode of the transistor exceeds the powersupply voltage Vps, the lowering speed of the voltage at the gateelectrode of the transistor is slowed and the lowering speed of thedrain current of the transistor is slowed. As a result, the suppressioncapability of the surge voltage becomes stable.

However, in such a case that the capacitor is utilized, the loweringspeed of the voltage at the gate electrode is not slowed, but also thelowering speed of the drain current of the transistor is not slowedunless the voltage of the drain electrode exceeds the power supplyvoltage. As a result, there are some possibilities that the surgevoltage can be hardly suppressed.

As a consequence, in accordance with this fourth embodiment, such aswitching circuit capable of improving the suppression capability of thesurge voltage can be provided by performing the below-mentioned manner.That is, not only the timing for slowing the changing speed of thevoltage at the control electrode of the transistor becomes stable, butalso the changing speed of the voltage at the control electrode of thetransistor is slowed at such a earlier timing than the timing when thevoltage at the major electrode of the transistor on the high voltageside exceeds the power supply voltage.

Now, a preferred semiconductor mode of this fourth embodiment will bedescribed. A switching circuit of the fourth embodiment utilizes afield-effect type transistor. The switching circuit of the fourthembodiment uses a MOSFET. Both a surge voltage suppressing circuit andthe transistor have been manufactured in the same semiconductorsubstrate.

FIG. 17 shows a circuit diagram of a switching circuit 210 equipped witha field-effect type transistor (n type MOSFET) 5. The transistor 5 hasbeen employed by series-connecting a power supply PS to a load R1between a drain electrode D and a source electrode S of this transistor5. A stray inductance “L” is present in a wiring line between thetransistor 5 and the load R1. The switching circuit 210 switchesturn-ON/OFF operations of the transistor 5 based upon a driving voltageVin outputted from the control circuit 3. The diving voltage Vin isinputted to the gate electrode G.

Since the switching circuit 210 switches turn-ON/OFF operations of thetransistor 5, the switching circuit 210 switches such a condition thatthe power supply voltage Vps supplied by the power supply PS is appliedto the load R1, and also, such a condition that the power supply voltageVps supplied by the power supply PS is not applied to the load R1.

It should also be noted that the drain electrode D corresponds to amajor electrode on the high voltage side, the source electrode Scorresponds to another major electrode on the low voltage side, and thegate electrode G corresponds to the control electrode.

The switching circuit 210 has been equipped with the transistor 5, acontrol circuit 3, and a surge voltage measure circuit 212. While thecontrol circuit 3 has been connected to the gate electrode G of thetransistor 5, the control circuit 3 outputs to the gate electrode G ofthe transistor 5 such a driving voltage Vin having a rectangular wave inwhich a voltage for turning ON the transistor 5 and another voltage forturning OFF the transistor 5 alternately appear. The surge voltagemeasure circuit 212 has been equipped with a series circuit 230 and avoltage adjusting circuit 220. One terminal of the series circuit 230has been connected to a first junction point 241 between the gateelectrode G of the transistor 5 and the control circuit 3. The otherterminal of the series circuit 230 has been connected to a secondjunction point 251 between the drain electrode D of the transistor 5 andthe load R1. The series circuit 230 has a first capacitor 7 a and afirst diode 9 a series-connected to the first capacitor 7 a. A cathodeof the first diode 9 a has been connected to the side of the firstjunction point 241, whereas an anode of the first diode 9 a has beenconnected to the side of the second junction point 251. The voltageadjusting circuit 220 has been connected to a third junction point 233of a connection line which connects the first capacitor 7 a to the firstdiode 9 a. The voltage adjusting circuit 220 lowers a voltage at thethird junction point 233 when the transistor 5 is turned ON, as comparedwith that when the transistor 5 is turned OFF. In other words, thevoltage adjusting circuit 220 may alternatively lower a charging voltageof the first capacitor 7 a when the transistor 5 is turned ON, ascompared with that when the transistor 5 is turned OFF. The surgevoltage measure circuit 212 containing the series circuit 230 has beenformed in the same semiconductor substrate as that of the transistor 5.

Referring now to FIG. 18A to FIG. 18C, a description is made ofoperations of the switching circuit 210. FIG. 18A to FIG. 18C showchanging patterns of voltages VD at the drain electrode D of thetransistor 5, namely represent time elapse variations of the voltage VDin transition time periods during which the transistor 5 is turned OFF.FIG. 18A is a case where the surge voltage measure circuit 212 is notprovided, and is a comparison example. FIG. 18B is a case where only theseries circuit 230 within the surge voltage measure circuit 212 isprovided, and is another comparison example. FIG. 18C is a case of theswitching circuit 210 according to this fourth embodiment.

Referring to FIG. 18A, a description is made of operations in such acase where the surge voltage measure circuit 212 is not provided.

In the case that the surge voltage measure circuit 212 is not provided,if the transistor 5 is turned OFF at timing “Toff”, then the voltage VDof the drain electrode D is increased. Then, in a final stage of atransition time period for turning OFF the transistor 5, a surge voltagehas been generated due to the drain current ID of the transistor 5 andthe stray capacitance L.

Next, a description is made of operations in such a case that only theseries circuit 230 within the surge voltage measure circuit 212 isprovided with reference to FIG. 18B.

Firstly, such a condition that the transistor 5 has been turned OFF andis under stable will now be considered. In the switching circuit of thiscomparison example, when the transistor 5 has been turned OFF and isunder stable condition, a voltage (Vps−V_(F)) is being applied to thefirst capacitor 7 a. This voltage (Vps−V_(F)) is obtained by subtractinga forward direction voltage V_(F) of the first diode 9 a from the powersupply voltage Vps. When the transistor 5 has been turned OFF and isunder stable condition, the first capacitor 7 a has been charged to thisvoltage (Vps−V_(F)). In other words, a voltage “V1” at the thirdjunction point 233 becomes (Vps−V_(F)).

Next, when the transistor 5 is turned ON, the voltage VD of the drainelectrode D is decreased. As a result, since a reverse bias voltage isapplied to the first diode 9 a, the first diode 9 a cannot bedischarged, but the charged voltage of the first capacitor 7 a ismaintained at the previously charged voltage (Vps−V_(F)) while thetransistor 5 is turned ON.

Next, when the transistor 5 is turned OFF at timing “Toff”, the voltageVD of the drain electrode D is increased. When the voltage VD of thedrain electrode D exceeds a total voltage(V1+V_(F)=Vps−V_(F)+V_(F)=Vps), namely the power supply voltage Vps, aforward direction voltage is applied to the first diode 9 a (timing“Tb”of FIG. 18B). The total voltage is defined by adding the voltage V1(Vps−V_(F)) of the third junction point 233 to the forward directionvoltage V_(F) of the first diode 9 a. When the forward direction voltageis applied to the first diode 9 a at the timing “Tb”, a charging currentstarts to flow through the first capacitor 7 a. Thereafter, the loweringspeed of the voltage at the gate electrode G of the transistor 5 isslowed, and the lowering speed of the drain current ID of the transistor5 is lowered. As a result, after the timing “Tb”, the steep increase ofthe drain voltage VD of the drain electrode D is suppressed, and thus,the surge voltage is suppressed to the low surge voltage.

However, if the voltage adjusting circuit 220 is not present, then sucha timing is fixed to the timing “Tb” at which the voltage of the drainelectrode D becomes equal to the power supply voltage Vps. As a result,there are some cases that the surge voltage cannot be sufficientlysuppressed. The above-described timing implies such a timing when thelowering speed of the voltage at the gate electrode G of the transistor5 is slowed, and the lowering speed of the drain current ID of thetransistor 5 is lowed, and also, the lowering speed of the voltage atthe drain electrode D of the transistor 5 is slowed.

Next, a description is made of operations when the voltage adjustingcircuit 220 has been provided with reference to FIG. 18C.

As explained in FIG. 18B, the timing at which the charge current startsto flow through the first capacitor 7 a corresponds to timing at which aforward direction voltage starts to be effected to the first diode 9 a.The timing at which the forward direction voltage starts to effect thefirst diode 9 a is such a timing when the voltage VD of the drainelectrode D exceeds a total voltage (V1+V_(F)) obtained by adding thevoltage V1 of the third junction point 233 and the forward directionvoltage V_(F) of the first diode 9 a. As a consequence, if the voltageV1 of the third junction point 233 is adjusted to a low voltage, thenthe timing at which the forward direction voltage starts to effect thefirst diode 9 a becomes early, and such a condition that the chargecurrent starts to flow through the first capacitor 7 a when the voltageVD of the drain electrode D is low can occur.

The voltage adjusting circuit 220 decreases the electron charges chargedin the first capacitor 7 a while the transistor 5 is turned OFF when thetransistor 5 is turned ON so as to decrease the voltage V1 of the thirdjunction point 233 up to a voltage “Vt” (refer to FIG. 18C). Thisvoltage “Vt” may have a predetermined magnitude, or may alternativelyhave different magnitudes every time the switching operation isrepeated. If the voltage V1 of the third junction point 233 when thetransistor 5 is turned ON is decreased up to the voltage Vt, asrepresented in FIG. 18C, the timing “Tc” at which the charge currentstarts to flow through the first capacitor 7 a becomes early. In otherwords, when the voltage VD of the drain electrode D is low, the chargecurrent starts to flow through the first capacitor 7 a; the loweringspeed of the voltage at the gate electrode G of the transistor 5 isslowed; the lowering speed of the drain current ID of the transistor 5is slowed; and the changing sped of the voltage at the drain electrode Dis slowed. That is to say, the charging current starts to flow throughthe first capacitor 7 a at such a timing which is sufficiently earlierthan the timing when the surge voltage becomes a peak voltage; thelowering speed of the voltage at the gate electrode G of the transistor5 is slowed; the lowering speed of the drain current ID of thetransistor 5 is slowed; and the changing speed of the voltage at thedrain electrode D is slowed. As a result, the increase of the surgevoltage can be considerably suppressed.

It should also be understood that while the first capacitor 7 a has beenarranged on the side of the drain electrode D of the transistor 5 andthe first diode 9 a has been arranged on the side of the gate electrodeG of the transistor 5, if the voltage adjusting circuit 220 isadditionally provided, then such a phenomenon is obtained. That is tosay, the charge current starts to flow through the first capacitor 7 aat such a timing which is sufficiently earlier than the timing when thesurge voltage becomes a peak voltage; the lowering speed of the voltageat the gate electrode G of the transistor 5 is slowed; and the changingspeed of the voltage at the drain electrode D is slowed. As a result,increasing of the surge voltage can be considerably suppressed.

FIG. 19 shows a concrete circuit diagram of the voltage adjustingcircuit 220.

The voltage adjusting circuit 220 has been connected between the thirdjunction point 233 and the anode of the first diode 9 a, while the thirdjunction point 233 connects the first capacitor 7 a to the first diode 9a. The voltage adjusting circuit 220 has been equipped with such acircuit that a second capacitor 7 b has been series-connected to thesecond diode 9 b. A cathode of the second diode 9 b has been connectedto the anode side of the first diode 9 a, and an anode of the seconddiode 9 b has been connected to the side of the third junction point233.

FIG. 20A to FIG. 20B show simulation results in the case that thetransistor 5 is driven by employing the switching circuit 210 of FIG.19. FIG. 20A shows time elapse changes as to a voltage VG at the gateelectrode G of the transistor 5, a voltage VD at the drain electrode Dthereof, and voltages V1, V2. It should be understood that the voltageV1 is a voltage at the third junction point 233, and the voltage V2corresponds to a voltage at a connection line between the second diode 9b and the second capacitor 7 b. FIG. 20B represents time elapse changesas to a drain current ID of the transistor 5, and currents I1 and I2.The current I1 is a current flowing through the first diode 9 a, whereasthe current I2 is a current flowing through the second diode 9 b. FIG.20A to FIG. 20B indicate the simulation results obtained when anelectrostatic capacitance C1 of the first capacitor 7 a is 1 nF; anelectrostatic capacitance C2 of the second capacitor 7 b is 0.5 nF; thepower supply voltage V_(PS) is 100 V; and the forward direction voltageV_(F) for the first diode 9 a and the second diode 9 b is 0.8 V.

Firstly, such a time period “T1” during which the transistor 5 has beenturned OFF and is under stable will now be considered. In the switchingcircuit 210, when the transistor 5 has been turned OFF and is understable condition, a voltage (Vps−V_(F)) is applied to the firstcapacitor 7 a and the second capacitor 7 b. This voltage (Vps−V_(F)) isobtained by subtracting a forward direction voltage V_(F) of the firstdiode 9 a from the power supply voltage Vps. A charge voltage of thefirst capacitor 7 a when the transistor 5 has been turned OFF is equalto (Vps−V_(F)), and the voltage V1 at the third junction point 233becomes (Vps−V_(F)). It should also be understood that the voltage V2between the second capacitor 7 b and the second diode 9 b becomes also(Vps−V_(F)). In other words, the second capacitor 7 b is not charged. Asshown in FIG. 20A to FIG. 20B, both the voltage V1 and the voltage V2 ofthe time period T1 are approximately 100 V.

Next, a time period “T2” during which the transistor 5 is turned ON willnow be considered. In the time period T2, firstly, the voltage VG of thegate electrode G of the transistor 5 is increased. Both voltage V1 andthe voltage V2 are increased in conjunction with increasing of the gatevoltage VG. When the voltage VG of the gate electrode G exceeds athreshold value, the drain current Id flows between the source electrodeS and the drain electrode D of the transistor 5, so that the transistor5 is turned ON. When the transistor 5 is turned ON, the voltage VD atthe drain electrode D is decreased. When the voltage VD of the drainelectrode D is decreased, a reverse bias voltage is applied to the firstdiode 9 a, so that the first diode 9 a is cut off. On the other hand, aforward direction voltage is applied to the second diode 9 b. When adifference between the voltage V2 and the voltage VD of the drainelectrode D exceeds the forward direction voltage V_(F) of the seconddiode 9 b, the second diode 9 b becomes conductive.

Referring now to FIG. 19, a phenomenon occurred at this time will bedescribed in more detail. When the transistor 5 is under OFF state, thefirst capacitor 7 a present between the third junction point 233 and thefirst junction point 241 has been charged. The third junction point 233corresponds to the high voltage side, whereas the first junction point241 corresponds to the low voltage side. At this time, the potential V1at the third junction point 233 is equal to the potential V2 at theconnection line between the second capacitor 7 b and the second diode 9b, and the charge voltage of the second capacitor 7 b is equal to zero.Next, when the transistor 5 is turned ON, as previously described, thesecond diode 9 b becomes conductive, and a portion of the electronswhich have been charged in the first capacitor 7 a is moved to thesecond capacitor 7 b so as to charge the second capacitor 7 b. At thistime, since the transistor 5 is under ON state, the second junctionpoint 251 is the lower voltage side with respect to the third junctionpoint 233. As a consequence, when the transistor 5 is turned ON, it ispossible to evaluate that a parallel circuit constituted of the firstcapacitor 7 a and the second capacitor 7 b is arranged. In other words,an electrostatic capacitance between the first junction point 241 andthe second junction point 251 becomes a series electrostatic capacitancemade of the first capacitance 7 a and the second capacitance 7 b. As aresult, the voltage V1 of the third junction point 233 is decreased. Inother words, a portion of the electrons charged in the first capacitor 7a moves to the second capacitor 7 b. As a result, the voltage V1 of thethird junction 233 is decreased. As indicated in FIG. 20B, in atransition time period during which the transistor 5 is turned, thecurrent I2 flows through the first capacitor 7 a, the second capacitor 7b, the second diode 9 b, and the second junction point 251, so that thevoltage V1 of the third junction point 233 is decreased. It can beunderstood that the decreasing amount of the voltage V1 at the thirdjunction point 233 can be adjusted by adjusting a ratio of theelectrostatic capacitance of the first capacitor 7 a to that of thesecond capacitor 7 b.

In a time period “T3” during which the transistor 5 has been completelyturned ON, both the first diode 9 a and the second diode 9 b are turnedOFF, so that both the voltage V1 and the voltage V2 are maintained.

Next, a description is made of a time period “T4” during which thetransistor 5 is turned OFF. When the transistor 5 is turned OFF, thevoltage VD of the drain electrode D is increased. If the voltage VD ofthe drain electrode D exceeds a total voltage (V1+V_(F)=Vt+V_(F))calculated by adding the charge voltage V1 charged in the firstcapacitor 7 a to the forward direction voltage V_(F) of the first diode9 a, then the first diode 9 a becomes conductive. When the first diode 9a becomes conductive, the current I1 starts to flow through the firstcapacitor 7 a; the lowering speed of the voltage at the gate electrode Gof the transistor 5 is slowed; and the lowering speed of the draincurrent ID of the transistor 5 is slowed. As a consequence, thereafter,the steep increase of the voltage VD of the drain electrode D can besuppressed, so that the surge voltage can be suppressed to a low surgevoltage.

Since these operations are repeatedly carried out, the switching circuit210 can drive the transistor 5 while suppressing the surge voltage.

FIG. 21 shows a variation as to the voltage VD of the drain electrode Dof the transistor 5 in the transition time period during which thetransistor 5 is turned OFF. In this drawing, a variation result obtainedwhen the electrostatic capacitance C1 of the first capacitor 7 a and theelectrostatic capacitance C2 of the second capacitor 7 b are changed isfurther indicated in combination with the above-described variation. Itshould also be noted that indication “WITHOUT MEASURE” implies avariation result obtained when the surge voltage measure circuit 212 isnot provided.

As represented in FIG. 21, when one case where the electrostaticcapacitance C1 of the first capacitor 7 a is 1 nF is compared withanother case where the electrostatic capacitance C1 thereof is 0.5 nF,it can be seen that inclinations of the voltages VD of the drainelectrode D are different from each other. The magnitude of theelectrostatic capacitance C1 of the first capacitance 7 a determines theinclination of the voltage VD of the drain electrode D.

As shown in FIG. 21, when one case where a ratio of the electrostaticcapacitance of the first capacitor 7 a to the electrostatic capacitanceof the second capacitor 7 b is 1:1 is compared with another case where aratio thereof is 1:0.5, timing at which the voltages VD of the drainelectrode D start to be inclined is different from each other. The ratioof the electrostatic capacitance of the first capacitor 7 a to that ofthe second capacitor 7 b determines the timing when the voltage VD ofthe drain electrode D starts to be inclined.

FIG. 22 shows a variation as to the drain current ID of the transistor 5in the transition time period during which the transistor 5 is turnedOFF.

As represented in FIG. 22, when one case where the electrostaticcapacitance C1 of the first capacitor 7 a is 1 nF is compared withanother case where the electrostatic capacitance C1 thereof is 0.5 nF,it can be seen that inclinations of the drain current ID of thetransistor 5 are different from each other. The magnitude of theelectrostatic capacitance C1 of the first capacitance 7 a determines theinclination of the drain current ID of the transistor 5.

As shown in FIG. 22, when one case where a ratio of the electrostaticcapacitance of the first capacitor 7 a to the electrostatic capacitanceof the second capacitor 7 b is 1:1 is compared with another case where aratio thereof is 1:0.5, timing at which the drain currents ID of thetransistor 5 start to be inclined differ from each other. The ratio ofthe electrostatic capacitance of the first capacitor 7 a to that of thesecond capacitor 7 b determines the timing when the drain current ID ofthe transistor 5 starts to be inclined.

As can be understood from the results of FIG. 21 and FIG. 22, since theelectrostatic capacitances of the first capacitor 7 a and the secondcapacitor 7 b are adjusted, the characteristics of the transition timeperiods during which the transistor 5 is turned OFF can be improved.

In particular, as represented in FIG. 21, the switching circuit 210 ofthe fourth embodiment can steeply vary the voltage VD of the drainelectrode D in the beginning stage of the transition time period duringwhich the transistor 5 is turned OFF, and can gently vary the voltage VDof the drain electrode D in the final stage of the transition timeperiod. As a consequence, since the voltage VD of the drain electrode Dis steeply varied in the beginning stage of the transition time period,increasing of the turn-OFF loss can be suppressed, whereas since thevoltage VD of the drain electrode D is gently varied in the final stageof the transition time period, increasing of the surge voltage can besuppressed.

FIG. 23 shows a trade-off curve between a turn-OFF loss and a surgevoltage, which is present in the case that the transistor 5 is driven. Abroken line of this drawing indicates a trade-off curve in case of“WITHOUT MEASURE.”

As apparent from FIG. 23, the switching circuit 210 of the fourthembodiment can reduce the surge voltage while the turn-OFF loss is notsubstantially increased. The switching circuit 210 of this fourthembodiment can overcome the trade-OFF relationship existed between theturn-OFF loss and the surge voltage.

The below-mentioned comparison examples indicate simulation resultsobtained in the case where as the electrostatic capacitances of thefirst capacitance 7 a and the second capacitance 7 b, extremeelectrostatic capacitances are employed. It should be noted that thearrangements of the switching circuits employed in the below-mentionedcomparison examples are identical to that of the switching circuit 210shown in FIG. 19.

Comparison Example 1

Results shown in FIG. 24A to FIG. 24B correspond to such a case that theelectrostatic capacitance C1 of the first capacitor 7 a is 1 nF, and theelectrostatic capacitance C2 of the second capacitor 7 b is 0.1 pF. Theelectrostatic capacitance C2 of the second capacitor 7 b is extremelysmall. As a consequence, the comparison example 1 is essentiallyequivalent to such a case that the second capacitor 7 b is not present.

As shown in FIG. 24A to FIG. 24B, in the comparison example 1, thecurrent I2 does not flow in the transition time period during which thetransistor 5 is turned ON. As a result, the voltage V1 at the thirdjunction point 233 is not decreased.

Comparison Example 2

Results shown in FIG. 25A to FIG. 25B correspond to such a case that theelectrostatic capacitance C1 of the first capacitor 7 a is 0.1 pF, andthe electrostatic capacitance C2 of the second capacitor 7 b is 1 nF.The electrostatic capacitance C1 of the first capacitor 7 a is extremelysmall.

As shown in FIG. 25A to FIG. 25B, in the comparison example 2, thecurrent I2 does not flow in the transition time period during which thetransistor 5 is turned ON. Furthermore, the current I1 does not flow inthe transition time period during which the transistor 5 is turned OFF.As a consequence, the comparison example 2 can be evaluated to beessentially equivalent to such a case that the surge voltage measurecircuit 212 is not provided.

Comparison Example 3

Results shown in FIG. 26A to FIG. 26B correspond to such a case that theelectrostatic capacitance C1 of the first capacitor 7 a is 1 nF, and theelectrostatic capacitance C2 of the second capacitor 7 b is 10 nF. Theelectrostatic capacitance C2 of the second capacitor 7 b is extremelylarge.

As shown in FIG. 26A to FIG. 26B, in the comparison example 3, theexcessive current I2 flows in the transition time period during whichthe transistor 5 is turned ON. As a result, the voltage V1 at the thirdjunction point 233 is excessively decreased. As a consequence, in thetransition time period during which the transistor 5 is turned OFF, thevoltage VD at the second junction point is inclined from the beginningstage thereof. In the switching circuit of the comparison example 3, theturn-OFF loss is increased.

As a substitution idea, for instance, in the switching circuit 210 ofFIG. 19, the series circuit constructed of the second capacitor 7 b andthe second diode 9 b, the parallel circuit constructed by the firstdiode 9 a, and the first capacitor 7 a may be alternatively arrangedbetween the first junction point 241 and the second junction point 251in an opposite positional relationship. Alternatively, the arrangingsequence between the second capacitor 7 b and the second diode 9 b maybe inversed.

Fifth Embodiment

As shown in FIG. 28, a switching circuit 300 according to a fifthembodiment of the present disclosure has been arranged by containing atransistor (MOSFET) 5, a load resistor R1, a gate control circuit 3, agate resistor 336, a power supply PS, a diode 9 a, and a capacitor 7.

The switching circuit 300 contains a MOSFET 5 corresponding to theswitching element. The MOSFET 5 is an N channel type MOSFET. A drainterminal D of the MOSFET 5 is connected via the load resistor R1 to apositive polarity side of a power supply PS. A source terminal S of theMOSFET 5, a negative polarity side of the power supply PS, and a groundterminal Y of the gate control circuit 3 are grounded. A gate terminal Gof the MOSFET 5 is connected via a gate resistor 336 to a controlvoltage output terminal X of the gate control circuit 3.

The switching circuit 300 further contains a surge voltage suppressingcircuit corresponding to a series circuit arranged by a diode 9 a and acapacitor 7. The surge voltage suppressing circuit connects a signalpath between the drain and the gate of the MOSFET 5. The drain terminalD of the MOSFET 5 is connected via the capacitance 7 to an anodeterminal of the diode 9 a. The gate terminal G of the MOSFET 5 isconnected to a cathode terminal of the diode 9 a.

The gate control circuit 3 outputs a control voltage (gate voltage) “VG”for turning ON/OFF the MOSFET 5 with respect to the gate terminal G ofthe MOSFET 5 from the control voltage output terminal X. When the gatevoltage VG is lower than a threshold voltage, the MOSFET 5 is under OFFstate. When the gate voltage VG is higher than, or equal to thethreshold voltage (for instance, 5V to 15V), the MOSFET 5 is under ONstate.

In the case that the MOSFET 5 is under OFF state, the drain voltage VDof the drain terminal D is maintained at the power supply voltage Vps.As a result, a forward direction bias is applied to the diode 9 a, andthus, is brought into a conduction state. As a consequence, a potentialdifference between the gate terminal G and the drain terminal D of theMOSFET 5 becomes equal to the power supply voltage Vps, and a chargevoltage Vc of the capacitor 7 becomes equal to such a voltage calculatedby subtracting the forward direction voltage V_(F) of the diode 9 a fromthe power supply voltage Vps. It should also be noted that since theforward direction voltage V_(F) of the diode 9 a is sufficiently low(for instance, approximately 0.7 V in silicon diode), as compared withthe normal power supply voltage, the charge voltage Vc between theterminals of the capacitor 7 becomes substantially equal to the powersupply voltage Vps.

Next, when the MOSFET 5 is brought into an ON state, the drain voltageVD of the drain terminal D is lowered from the power supply voltage Vpsup to a voltage “Vds” between the source and the drain (approximately,several V). At this time, the charge voltage Vc of the capacitor 7 iskept at the power supply voltage Vps, and a reverse direction bias isapplied to the diode 9 a, so that the diode 9 a is brought into a cutoff state.

Next, if the gate voltage VG is lowered, so that the ON state of theMOSFET 5 is transferred to an OFF state, then the drain voltage VD isincreased. As shown in FIG. 29, since a stray inductance “L” iscontained in the switching circuit 300, a surge voltage is generated dueto this stray inductance “L”, so that the drain voltage VD is increasedhigher than, or equal to the power supply voltage Vps. In connectionwith the increase of the drain voltage VD, if the voltage Vdg betweenthe gate and the drain becomes higher than, or equal to a summationvoltage between the voltage Vc held in the capacitor 7 and the forwarddirection voltage V_(F) of the diode 9 a, then the diode 9 a is biasedalong the forward direction to become conductive. Thus, a current flowsthrough the gate resistor 336, so that the voltage at the gate terminalis increased. As a result, the current Id between the source and thedrain of the MOSFET 5 is increased, and a lowering speed of the currentId between the source and the drain is slowed, so that the drain voltageVD is lowered.

As previously described, since the increase of the drain voltage VD isnegatively fed back to the gate terminal via the series circuitconstructed of the diode 9 a and the capacitor 7, the current Id betweenthe source and the drain of the MOSFET 5 is increased, so thatincreasing of the drain voltage VD caused by the surge voltage can besuppressed.

It should be noted that the above-described switching circuit 300 may bealternatively applied to an inverter circuit, a DC-to-DC convertercircuit, and the like.

A description is made of an embodiment as to the switching circuit 300equipped with the N channel type MOSFET 5. In this case, a suppressioneffect of a surge voltage is indicated in such a case that the capacitor7 is selected to be 1 nF, and also be 10 nF, while the power supplyvoltage Vps is selected to be 100 V.

In the case that the MOSFET 5 is under OFF state, the diode 9 a isbrought into a conduction state, so that a potential difference betweenthe gate G and the drain D of the MOSFET 5 becomes equal to the powersupply voltage Vps. A voltage Vc obtained by subtracting the forwarddirection voltage V_(F) (approximately 0.7 V) of the diode 9 a from thepower supply voltage Vps (=100 V) is held in the capacitor 7. When theMOSFET 5 becomes an ON state, the drain voltage VD is lowered to severalvolts, and the capacitor 7 maintains approximately 100 V, and a reversedirection bias is applied to the diode 9 a, so that the diode 9 a isbrought into a cut off state. If the control voltage VG is lowered, sothat the ON state of the MOSFET 5 is changed into an OFF state, then thedrain voltage VD is increased.

FIG. 30 represents a temporal change of a current flowing through thediode 9 a when the MOSFET 5 is brought into the OFF state and a surgevoltage is generated at the drain terminal D. Also, FIG. 31 shows atemporal change of the gate voltage VG which is applied to the gateterminal G when the surge voltage is generated at the drain terminal D.Further, a temporal change of the gate voltage in the case where thesurge voltage suppressing circuit is not provided is indicated in FIG.31 as a comparison example.

If a surge voltage is generated during a time period from a time instantof 1.57×10⁻⁶ seconds to a time instant of 1.60×10⁻⁶ seconds, and thevoltage Vdg between the gate and the drain of the MOSFET 5 becomeshigher than, or equal to a summation voltage between the voltage Vc heldin the capacitor 7 and the forward direction voltage V_(F) of the diode9 a, namely the power supply voltage Vps of 100V, then the diode 9 a isbiased in the forward direction, so that the diode 9 a becomesconductive. As a result, a current flows through the gate resistor 336and thus the voltage VG of the gate terminal G is increased. At thistime, as shown in FIG. 30, a circuit impedance in the case that thecapacitance value of the capacitor 7 of the surge voltage suppressingcircuit is 10 nF (indicated by line “XXXB”) becomes smaller than acircuit impedance in the case that the capacitance value of thecapacitor 7 of the surge voltage suppressing circuit is 1 nF (indicatedby line “XXXA”), so that a larger current flows. As a result, asrepresented in FIG. 31, such a gate voltage VG in the case that thecapacitance value of the capacitor 7 of the surge voltage suppressingcircuit is 10 nF (indicated by line “XXXIB”) which is lower than a gatevoltage VG in the case that the capacitance value of the capacitor 7 ofthe surge voltage suppressing circuit is 1 nF (indicated by line“XXXIA”) is applied to the gate terminal G of the MOSFET 5. It shouldalso be noted that when the surge voltage suppressing circuit is notprovided, no current flows through the gate resistor 336, and thus, thegate voltage VG is not increased.

FIG. 32 indicates a temporal change, of the drain voltage VD when theMOSFET 5 is turned OFF. Since the gate voltage VG is increased, if thecurrent Id between the source and the drain of the MOSFET 5 isincreased, then the drain voltage VD is decreased, so that the adverseinfluence of the surge voltage is suppressed. At this time, as shown inFIG. 32, the suppression effect of the surge voltage achieved in thecase that the capacitance value of the capacitor 7 of the surge voltagesuppressing circuit is 10 nF (indicated by line “XXXIIB”) becomes higherthan the suppression effect of the surge voltage achieved in the casethat the capacitance value of the capacitor 7 of the surge voltagesuppressing circuit is 1 nF (indicated by line “XXXIIA”).

It should also be understood that when the surge voltage suppressingcircuit is not employed, no current flows through the gate resistor 336,and thus, the gate voltage VG is not increased.

In such a case that a suppressing circuit for a surge voltage is notprovided, as represented in FIG. 33, there is a trade-off relationshipbetween a surge voltage and a loss produced when the MOSFET 5 is turnedOFF. In this graph, a product made by a drain voltage VD and a draincurrent Id is integrate for a turn-OFF time period is defined as a lossoccurred when the MOSFET 5 is turned OFF. In other words, in order toreduce the loss occurred when the MOSFET 5 is turned OFF, a turn-OFFtime must be shortened. However, since the turn-OFF time is shortened, asurge voltage caused by a stray inductance is increased. On the otherhand, if the surge voltage suppressing circuit of this fifth embodimentis employed, as indicated in FIG. 33, then the surge voltage can bereduced while the loss occurred when the MOSFET 6 is turned OFF issuppressed to a small loss. At this time, the suppression effect of thesurge voltage achieved in the case that the capacitance value of thecapacitor 7 of the surge voltage suppressing circuit is 10 nF (indicatedby line “XXXIIIB”) becomes higher than the suppression effect of thesurge voltage achieved in the case that the capacitance value of thecapacitor 7 of the surge voltage suppressing circuit is 1 nF (indicatedby line “XXXIIIA”). In this case, a dot line “XXXIIIC” indicates atrade-off curve between a surge voltage and a turn-OFF loss of ageneral-purpose power MOSFET.

As indicated in FIG. 36A to FIG. 45, a portion of the switching circuit300 may be constituted by forming the MOSFET 5, the diode 9 a, and thecapacitor 7 on the same semiconductor substrate 20. Any of the examplesshown in FIG. 36A to FIG. 45 corresponds to such a semiconductor modethat the MOSFET 5 and the capacitor 7 have been formed in thesemiconductor substrate 20, and the diode 9 a has been formed on thesemiconductor substrate 20. The MOSFET 5, the diode 9 a, and thecapacitor 7 have been arranged in an integral body by utilizing thesemiconductor substrate 20, and have been formed as a single chip. Itshould also be noted that single crystalline silicon has been employedin the semiconductor substrate 20.

The semiconductor mode shown in FIG. 36A to FIG. 36B is such an examplethat the series circuit made of the diode 9 a and the capacitor 7 hasbeen connected between the drain D and the gate G of the MOSFET 5, andthe capacitor 7 has been inserted between the drain D of the MOSFET 5and the diode 9 a. FIG. 36A schematically indicates a sectional view ofa major portion of the semiconductor substrate 20. FIG. 36B shows anequivalent circuit of this semiconductor structure.

FIG. 36A indicates a vicinity of a boundary between an element regionand a periphery region. The element region corresponds to such a regionthat a semiconductor region group has been formed which is required inorder to temporally switch a signal path between the major electrodes(namely, drain D and source S) under conductive state and undernon-conductive state. The element region has been arranged at a centerside of the semiconductor substrate 20. The periphery region has beenarranged in such a manner that a peripheral portion of a central regionis surrounded. A terminal region has been formed in the peripheryregion. The terminal region has been arranged in such a manner that theperipheral portion of the central region is surrounded. The terminalregion corresponds to such a region that when the MOSFET 5 is under OFFstate, a voltage applied to the MOSFET 5 is loaded along a lateraldirection by expanding a depletion layer from the element region towarda side direction. It should be understood that widths of the terminalregion along the lateral direction indicated in FIG. 36A is representedin a compression mode for the sake of clear illustrations.

The semiconductor substrate 20 has been provided with an n⁺ type drainregion 22 and an n⁻ type drift region 24. The n⁺ type drain region 22has been formed on a rear surface of the semiconductor substrate 20,whereas the n⁻ type drift region 24 has been provided on the drainregion 22. Both the drain region 22 and the drift region 24 have beencontinuously provided along the lateral direction from the elementregion to the periphery region within the semiconductor substrate 20.

A p type body region 76, an n⁺ type source region 72, and a p⁺ type bodycontact region 74 have been provided on a front surface of thesemiconductor substrate 20 of the element region. The body region 76 hasbeen continuously provided on the front surface of the semiconductorsubstrate 20 of the element region along the lateral direction. Thesource region 72 and the body contact region 74 have been selectivelyprovided on the front surface of the semiconductor substrate 20. Thesource region 72 has been isolated from the drift region 24 by the bodyregion 76. The body contact region 74 contains higher dopantconcentration than that of the body region 76. Both the source region 72and the body contact region 74 have been electrically connected to thesource S provided on the semiconductor substrate 20.

A trench gate electrode 64 and a gate insulating film 62 have beenfurther formed on the front surface of the semiconductor substrate 20 ofthe element region. The trench electrode 64 has been extended from thefront surface of the semiconductor substrate 20 toward the rear surfacethereof, and has penetrated through the body region 76 and then hasreached the drift region 24. The gate insulating film 62 has covered thetrench gate electrode 64 so as to electrically insulate the trench gateelectrode 64 from the semiconductor substrate 20. The trench gateelectrode 64 is located opposite to the body region 76 via the gateinsulating film 62, while the body region 76 has isolated the sourceregion 72 from the drift region 24. Poly crystalline silicon has beenemployed in the trench gate electrode 64. A silicon oxide has beenemployed in the gate insulating film 62.

A selective oxide film 42 (one example of insulating film) has beenprovided on the front surface of the semiconductor substrate 20 of theperiphery region. A silicon oxide has been employed in the selectiveoxide film 42. The selective oxide film 42 can relax an electric fieldof the front surface of the semiconductor substrate 20 of the terminalregion. Generally speaking, there are many possibilities that suchstructures as, for example, a RESURF layer and a guard ring have beenprovided within the semiconductor substrate 20 located under theselective oxide film 42. These structures have been provided in order torelax electric field concentration occurred in the peripheral portion ofthe element region, and thus, so as to improve a withstanding voltage ofthe MOSFET 5. In FIG. 36A, for the sake of clear illustration, thesevery fine structures have been omitted.

An n type diffusion semiconductor region 38 and an n⁺ type surfacediffusion semiconductor region 32 have been provided on a front surfaceof the semiconductor substrate 20 which is located in a further sidedirection than the terminal region. Both the n type diffusionsemiconductor region 38 and the n⁺ type surface diffusion semiconductor32 have been provided on at least a portion of the front surface of thesemiconductor substrate 20 located in the further side direction thanthe terminal region. In other words, the n type diffusion semiconductorregion 38 and the n+ type surface diffusion semiconductor region 32 neednot be provided in such a manner that the peripheral portion of theterminal region is surrounded by these regions 32 and 38.

An embedded conductive region 36 and a coating insulation region 34 havebeen furthermore provided in a portion of the front surface of thesemiconductor substrate 20 which is located in a further side directionthan the terminal region. The embedded conductive region 36 has beenextended from the front surface of the semiconductor substrate 20 to therear surface thereof, and then has been stayed within the n typediffusion semiconductor region 38. The coating insulation region 34 hascovered the embedded conductive region 36 so as to electrically insulatethe embedded conductive region 36 from the semiconductor substrate 20.Poly crystalline silicon has been employed in the embedded conductiveregion 36. A silicon oxide has been employed in the coating insulationregion 34.

The embedded conductive region 36 has been located opposite to the ntype diffusion semiconductor region 38 via the coating insulation region34. As a consequence, the capacitor 7 has been constituted by theembedded conductive region 36, the coating insulation region 34, and then type diffusion semiconductor region 38. It should also be noted thatthe n type diffusion semiconductor region 38 has been formed in order tosuppress that a depletion layer is produced in a boundary between thecoating insulation region 34 and the n type diffusion semiconductorregion 38, and thus, so as to stabilize the capacitance of the capacitor7.

The embedded conductive region 36 has the same depth as that of thetrench gate electrode 64 of the element region. The coating insulationregion 34 has the same thickness as that of the gate insulating film 62of the element region. As a consequence, both the embedded conductiveregion 36 and the coating insulation region 34 can be formed at the sametime by utilizing the process for manufacturing the trench gateelectrode 64 of the element region and the gate insulating film 62.

The diode 9 a has been formed on the front surface of the selectiveregion 42. The diode 9 a has been provided with an anode semiconductorregion 54 containing a p type dopant, and a cathode semiconductor region52 containing an n type dopant. While the anode semiconductor region 54and the cathode semiconductor region 52 have been provided on the frontsurface of the selective oxide film 42, the anode semiconductor region54 and the cathode semiconductor region 52 are directly contactedthereto. As materials of the anode semiconductor region 54 and thecathode semiconductor region 52, poly crystalline silicon has beenemployed.

As represented in FIG. 36, the cathode semiconductor region 52 of thediode 9 a has been electrically connected to the trench gate electrode64 of the MOSFET 5. The anode semiconductor region 54 of the diode 9 ahas been electrically connected to the embedded conductive region 36 ofthe capacitor 7. In other words, the anode semiconductor region 54 ofthe diode 9 a has been electrically connected to one end of thecapacitor 7. The n type diffusion semiconductor region 38 has beenelectrically connected via the n⁺ type surface diffusion semiconductorregion 32 to the drain D. Also, the n type diffusion semiconductorregion 38 has been electrically connected via the drift region 24 andthe drain region 22 to the drain D. As a consequence, the other end ofthe capacitor 7 has been electrically connected to the drain D of theMOSFET 5. A switching circuit-purpose chip for embodying an equivalentcircuit shown in FIG. 36B has been arranged by electrically connectingthese electronic elements. It should also be understood that theseelectric connections may be constituted by, for example, aluminum wiringlines.

As previously described, if the diode 9 a and the capacitor 7 areconstructed on the same semiconductor substrate 20 as the MOSFET 5, thenthe diode 9 a and the capacitor 7 need not be separately prepared. As aconsequence, a semiconductor chip for the switching circuit 300 can beconstructed without increasing a total number of these components. Ifthe MOSFET 5, the diode 9 a, and the capacitor 7 have been provided onthe same semiconductor substrate 20, then the switching circuit 300 canbe arranged by employing a small number of the components. Thus, theswitching circuit 300 can be furthermore made compact.

FIG. 37 and FIG. 38 indicate a technical idea for changing a capacitanceof the capacitor 7. The capacitance of the capacitor 7 is adjusted basedupon a thickness and an area of the coating insulation region 34. Thecapacitance of the capacitor 7 is inverse proportion to the thickness ofthe coating insulation region 34, and is direct proportion to the areaof the coating insulation region 34.

As indicated in FIG. 37, if the embedded conductive region 36 and thecoating insulation region 34 are formed up to a deep position of thesemiconductor substrate 20 so as to increase the area of the coatinginsulation region 34, then the capacitance of the capacitor 7 isincreased. On the other hand, as shown in FIG. 38, if the thickness ofthe coating insulation region 34 is increased, then the capacitance ofthe capacitor 7 can be decreased. As previously explained, thesemiconductor modes as to the embedded conductive region 36 and thecoating insulation region 34 are contrived, so that the capacitance ofthe capacitor 7 can be readily adjusted. As apparent from the foregoingdescription, the thickness of the covering insulator 34 may be madethicker, and further, both the embedded conductive region 36 and thecoating insulation region 34 may be alternatively extended up to adeeper position of the semiconductor substrate 20.

FIG. 39 indicates such a semiconductor structural example that a superjunction structure has been formed in the drift region 24 of the elementregion. The drift region 24 of the super junction structure has beenformed by a repetition structure made of an n type column 24 acontaining an n type dopant, and a p type column 24 b containing a ptype dopant. The impurity concentration of the p type column 24 b issubstantially equal to that of the body region 76. In the super junctionstructure, when the MOSFET 5 is turned OFF, a depletion layer isexpanded from the pn junction between the n type column 24 a and the ptype column 24 b, and thus, the drift region 24 can be substantiallycompletely depleted. As a result, if a pitch width between the n typecolumn 24 a and the p type column 24 b is adjusted, even when theimpurity concentration is increased, then the drift region 24 can besubstantially perfectly depleted. As a consequence, since the impurityconcentration of the n type column 24 a and the p type column 24 b canbe increased, the super junction structure is useful as such a techniquecapable of solving the trade-off relationship existed between theON-resistance (otherwise, ON voltage) of the MOSFET 5 and thewithstanding voltage.

Generally speaking, a super junction structure is formed by utilizingthe semiconductor substrate 20 which contains an n type dopant inrelatively high concentration. Concretely speaking, a plurality oftrenches are formed from the front surface of the semiconductorsubstrate 20 by performing an anisotropic etching process, and then, ptype columns 24 b are crystal-grown within these trenches. The remainingportions when the plural trenches are formed become n type columns 24 a.As a result, such a super junction structure that the n type columns 24a and the p type columns 24 b are repeatedly arranged may bemanufactured.

In the case where the super junction structure is employed in the driftregion 24 of the element region, the semiconductor substrate 20 isemployed which contains the n type dopant in relatively highconcentration. As a consequence, in the case where the super junctionstructure is employed in the drift region 24 of the element region, theimpurity concentration of the drift region 24 of the periphery region isincreased. Accordingly, in this example, even if the n type diffusionsemiconductor region 38 is not provided which was formed in the case ofFIG. 36, then it is possible to suppress such a phenomenon that thedepletion layer is produced at the boundary of the coating insulationregion 34.

As previously described, the technique for forming the MOSFET 5, thediode 9 a, and the capacitor 7 on the same semiconductor substrate 20may be applied to various sorts of transistors. Generally speaking,there are many possibilities that a periphery region of a transistor hasbeen provided with the selective oxide film 42. If the diode 9 a isprovided on the front surface of this selective oxide film 42, then thediode 9 a may be provided above the semiconductor substrate 20 of theperiphery region without any limitation to the sort of transistor. Sincethe space of the surface of the selective oxide film 42 is utilized, thediode 9 a can be easily formed in the semiconductor substrate 20 in anintegral body while increasing of the element area is suppressed. Also,if both the embedded conductive region 36 and the coating insulationregion 34 are utilized, then the capacitor 7 may be easily formed withinthe semiconductor region 20 of the periphery region. Since both theembedded conductive region 36 and the coating insulation region 34 havethe trench shapes, the capacitor 7 can be easily formed in thesemiconductor substrate 20 in an integral body while increasing of theelement area is suppressed.

FIG. 40 is an example in which the diode 9 a is not provided on thefront surface of the selective oxide film 42, but is formed to beembedded in the semiconductor substrate 20. In this example, the diode 9a has been arranged by arranging both an n⁺ type cathode semiconductorregion 350 containing an n type dopant and a p⁺ type anode semiconductorregion 352 containing a p type dopant on the front surface of the p typebody region 76 which is elongated along the side direction of the MOSFET5. The n⁺ type cathode semiconductor region 350 and the p⁺ type anodesemiconductor region 352 may be formed by diffusing the n type dopantand the p type dopant on the front surface of the semiconductorsubstrate 20 respectively by executing a thermal diffusion method, andthe like. As a consequence, the n⁺ type cathode semiconductor region 350and the p⁺ type anode semiconductor region 352 may be made of singlecrystalline silicon layers.

As shown in FIG. 40, the n⁺ type cathode semiconductor region 350 of thediode 9 a has been electrically connected to the trench gate electrode64 of the MOSFET 5. The p⁺ type anode semiconductor region 352 of thediode 9 a has been electrically connected to the embedded conductiveregion 36 of the capacitor 7. In other words, the p⁺ type anodesemiconductor region 352 of the diode 9 a has been electricallyconnected to one end of the capacitor 7. It should also be understoodthat these electric connections may be constituted by, for example,aluminum wiring lines.

A selective oxide film 42 (one example of insulating film) has beenprovided on the front surface of the semiconductor substrate 20 alongthe side direction of the p⁺ type anode semiconductor region 352. Asilicon oxide has been employed in the selective oxide film 42. Theselective oxide film 42 can relax an electric field of the front surfaceof the semiconductor substrate 20. Generally speaking, there are manypossibilities that such structures as, for example, a RESURF layer and aguard ring have been provided within the semiconductor substrate 20located under the selective oxide film 42. In FIG. 40, for the sake ofclear illustration, these very fine structures have been omitted.

FIG. 41 shows such an example that the position for the embedded regionof the diode 9 a and the position for the element separation region bythe selective oxide film 42 shown in FIG. 40 are replaced with eachother. In this example, the selective oxide film 42 has been formed onthe front surface of the semiconductor substrate 20 along the sidedirection of the element region of the MOSFET 5. A p type well region356 formed by diffusing a p type dopant by performing a thermaldiffusion method, and the like has been provided in a further sidedirection of this selective oxide film 42. The diode 9 a has beenarranged by arranging both an n⁺ type cathode semiconductor region 350containing an n type dopant and a p⁺ type anode semiconductor region 352containing a p type dopant on the front surface of this p type wellregion 356. The n⁺ type cathode semiconductor region 350 and the p⁺ typeanode semiconductor region 352 may be formed by diffusing the n typedopant and the p type dopant on the front surface of the semiconductorsubstrate 20 respectively by executing a thermal diffusion method, andthe like.

Also, similar to FIG. 40, the n⁺ type cathode semiconductor region 350of the diode 9 a has been electrically connected to the trench gateelectrode 64 of the MOSFET 5. The p⁺ type anode semiconductor region 352of the diode 9 a has been electrically connected to the embeddedconductive region 36 of the capacitor 7. In other words, the p⁺ typeanode semiconductor region 352 of the diode 9 a has been electricallyconnected to one end of the capacitor 7.

FIG. 42A to FIG. 42B indicate such a structural example that theposition of the diode 9 a and the position of the capacitor 7 shown inFIG. 40 are replaced with each other. In this example, the selectiveoxide film 42 has been provided on the surface of the semiconductorsubstrate 20 along the side direction of the element region of theMOSFET 5. Alternatively, a potential control layer 358 containing a ptype dopant may be provided on the front surface of the body region 76along the side direction of the element region of the MOSFET 5.

A capacitor 7 made of the embedded conductive region 36, the diffusionsemiconductor region 38, and the coating insulation region 34 has beenformed on the front surface of the semiconductor substrate 20 along theside direction of the selective oxide film 42. The capacitor 7 may bemanufactured in a similar manner to other examples subsequent to FIG.36.

Also, a p type well region 356 formed by diffusing a p type dopant byperforming a thermal diffusion method, and the like has been provided onthe front surface of the semiconductor substrate 20 along a further sideof the capacitor 7. A diode 9 a has been arranged by arranging both ann⁺ type cathode semiconductor region 350 containing an n type dopant anda p⁺ type cathode semiconductor region 352 containing a p type dopant onthe front surface of this p type well region 356. The n⁺ type cathodesemiconductor region 350 and the p⁺ type anode semiconductor region 352may be formed by diffusing the n type dopant and the p type dopant onthe front surface of the semiconductor substrate 20 respectively byexecuting a thermal diffusion method, and the like.

In the structure of FIG. 42A, the trench gate electrode 64 of the MOSFET5 has been electrically connected to the embedded conductive region 36of the capacitor 7. The cathode semiconductor region 350 of the diode 9a has been electrically connected to the surface diffusion semiconductorregion 32 of the capacitor 7. The anode semiconductor region 352 of thediode 9 a has been electrically connected to the drain D. In otherwords, a structural mode shown in FIG. 42B is such an arrangement thatthe capacitor 9 a and the capacitor 7 are replaced with each other inthe equivalent circuit of FIG. 36B. That is to say, as indicated in FIG.42B, this structural mode may be expressed by such an equivalent circuitthat the anode of the diode 9 a is connected to the drain D of theMOSFET 5, and the gate of the MOSFET 5 is connected via the capacitor 7to the cathode of the diode 9 a.

FIG. 43 represents a structural mode in which the connection between thetrench gate electrode 64 of the MOSFET 5 and the embedded conductiveregion 36 of the capacitor 7, and the connection between the cathodesemiconductor region 350 of the diode 9 a and the surface diffusionsemiconductor region 32 of the capacitor 7 are replaced with each other,which are employed in FIG. 42. That is to say, the trench gate electrode64 of the MOSFET 5 has been electrically connected to the surfacediffusion semiconductor region 32 of the capacitor 7. The cathodesemiconductor region 350 of the diode 9 a has been electricallyconnected to the embedded conductive region 36 of the capacitor 7. Theanode semiconductor region 352 of the diode 9 a has been electricallyconnected to the drain D. Also, the structural mode shown in FIG. 43 issimilarly expressed by the equivalent circuit indicated in FIG. 42B.

It should be understood that also other arrangements may bealternatively realized by replacing the connections among the embeddedconductive region 36 and the surface diffusion semiconductor region 32of the capacitor 7, and other structural portions. The connections maybe alternatively changed due to conditions as to wiring patterns formedon the surface of the semiconductor substrate 20.

FIG. 44 indicates an example in which a super junction structure hasbeen applied to the drift region 24 of the element region in FIG. 41.Also, FIG. 45 indicates an example in which a super junction structurehas been applied to the drift region 24 of the element region in FIG.40. As shown in FIG. 39, the drift region 24 of the super junctionstructure has been formed by a repetition structure made of an n typecolumn 24 a containing an n type dopant, and a p type column 24 bcontaining a p type dopant. The drift region 24 having the superjunction structure may be formed in a similar manner to that of FIG. 39.Since the drift region 24 is formed as the super junction structure, theimpurity concentration of the n type column 24 a and the p type column24 b can be increased, so that the super junction structure is useful assuch a technique capable of overcoming the trade-off relationshipexisted between the ON-resistance (otherwise, ON voltage) of the MOSFET5 and the withstanding voltage.

As shown in FIG. 40 to FIG. 45, even when such a structure that thediode 9 a is embedded in the semiconductor substrate 20 is employed, thediode 9 a may be formed in the same semiconductor substrate 20 as thecapacitor 7, or the MOSFET 5. As a result, the diode 9 a and thecapacitor 7 need not be separately prepared. As a consequence, thesemiconductor chip for the switching circuit 300 may be constructedwithout increasing the total number of structural components. Moreover,if the diode 9 a, the capacitor 7, and the MOSFET 5 are manufactured inthe same semiconductor substrate 20, then the switching circuit 300 maybe constructed by employing a smaller number of these structuralcomponents, and thus, may be made compact.

When the diode 9 a is formed on the selective oxide film 42, an areaefficiency used on the surface of the semiconductor substrate 20 may beincreased, whereas when the diode 9 a is embedded, the diode 9 a can beformed by single crystalline silicon, so that the rectificationcharacteristic may become superior.

Although the transistor has been described as the MOSFET in theabove-explained concrete example, even if the above-described MOSFET isreplaced by a bipolar transistor, or an IGBT, then similar operationsand effects may be achieved. In this alternative case, when thetransistor is replaced by the bipolar transistor, the transistor isarranged in such a manner that the gate G is replaced by a base “B”, thedrain D is replaced by a collector “C”, and the source S is replaced byan emitter “E.”

Sixth Embodiment

Preferred features of a driving circuit according to a sixth embodimentof the present disclosure will now be described. The driving circuit ofthis sixth embodiment drives a field-effect type transistor. A variableresistor has been formed in the same semiconductor substrate as that ofa transistor. The variable resistor is a pinch resistor. The pinchresistor has a structure in which a p type semiconductor region issandwiched by an n type semiconductor region. When a DC voltage which isapplied to the transistor is applied to the n type semiconductor region,the p type semiconductor region is essentially completely depleted. Thevariable resistor is an MOS (Metal Oxide Semiconductor) type resistor.The MOS type resistor has a stacked structure constructed of a conductorregion, an insulator region, and a p type semiconductor region. When theDC voltage which is applied to the transistor is applied to the n typesemiconductor region, the p type semiconductor region is essentiallycompletely depleted.

FIG. 46 shows a circuit diagram of a driving circuit 410 for driving afield-effect type transistor (n type MOSFET) 5. The transistor 5 hasbeen connected between a load R1 and the ground. A stray inductance of awiring line has been connected between the transistor 5 and the load R1.The driving circuit 410 applies a driving voltage “Vin” having arectangular waveform to a gate electrode G of the transistor 5, andswitches turning-ON/OFF operations of the transistor 5 based upon thisdriving voltage “Vin.” The driving circuit 410 switches theturning-ON/OFF operations of the transistor 5 so as to switch suchconditions that a DC voltage Vps of a voltage supply source PS isapplied, and is not applied to the load R1. A protection-purpose zenerdiode 422 has been connected between a source electrode “S” and the gateelectrode “G” of the transistor 5 in order to avoid that a voltagehigher than, or equal to a predetermined voltage is not applied to thegate electrode G.

The driving circuit 410 has been equipped with a driving voltagegenerating circuit 411, a fixed resistor R10, a first diode D10, asecond diode D12, and a variable resistor R12. The second diode D12 maybe alternatively deleted, if necessary. The fixed resistor R10, thefirst diode D10, the variable resistor R12, and the protection-purposezener diode 422 have been formed on the same semiconductor substrate asthat of the transistor 5. A concrete semiconductor mode will bedescribed in the below-mentioned embodiment.

A series circuit constructed of the fixed resistor R10 and the firstdiode D10 has been connected between the gate electrode G and a gateterminal G10. Another series circuit constructed of the variableresistor R12 and the second diode D12 has been connected between thegate electrode G and the gate terminal G10. The driving voltagegenerating circuit 411 has been electrically connected to the gateterminal G10. In other words, both the series circuit constructed of thefixed resistor R10 and the first diode D10 and another series circuitconstructed of the variable resistor R12 and the second diode D12 haveconstituted a parallel circuit between the driving voltage generatingcircuit 411 and the transistor 5. An anode of the first diode D10 hasbeen connected via the fixed resistor R10 to the gate terminal G10, anda cathode of the first diode D10 has been connected to the gateelectrode G of the transistor 5. An anode of the second diode D12 hasbeen connected via the variable resistor R12 to the gate electrode G ofthe transistor 5, and a cathode of the second diode D12 has beenconnected to the gate terminal G10.

A resistance value of the variable resistor R12 is adjusted based upon asource-to-drain voltage “Vds” of the transistor 5 which is measured by adrain voltage detecting means 450. The resistance value of the variableresistor R12 is adjusted to become small when the source-to-drainvoltage Vds of the transistor 5 is low. The resistance value of thevariable resistor R12 is adjusted to become large when thesource-to-drain voltage Vds of the transistor 5 is high.

FIG. 47A to FIG. 47E show operating waveform diagrams of the transistor5.

Firstly, a description is made of a transition time period during whichthe transistor 5 is turned ON. Since the second diode D12 has beenprovided along a reverse direction, the driving voltage Vin is suppliedto a wiring line on the side of the fixed resistor R10. When a signallevel of the driving voltage Vin is changed from a low level to a highlevel, the driving voltage Vin is converted into a positive gate current“Ig(+)” at the fixed resistor R10, and then, the positive gatecurrent“Ig(+)” is supplied to the gate electrode G of the transistor 5.When the positive gate current “Ig(+)” is supplied to the gate electrodeG of the transistor 5, electron charges are stored in the gate electrodeG. When the electron charges are stored in the gate electrode G, agate-to-source voltage “Vgs” of the transistor 5 is increased. When thegate-to-source voltage “Vgs” of the transistor 5 is increased, a draincurrent “Id” starts to flow from the drain electrode D of the transistor5 toward the source electrode S, so that the drain-to-source voltage Vdsis decreased. The state of the transistor 5 is transferred from the OFFstate to the ON state by performing these operation steps.

Next, a description is made of a transition time period “T10” duringwhich the transistor 5 is turned ON. When the signal level of thedriving voltage Vin becomes a low level from the high level, theelectron charges which have been stored in the gate electrode G aredischarged. Since the first diode D10 has been provided along thereverse direction, a negative gate current “Ig(−)” produced inconnection with the discharge of the electron charges flows toward thewiring line on the side of the variable resistor R12. In the beginningstage of the transition time period T10 during which the transistor 5 isturned OFF, since the drain-to-source voltage Vds is low, the resistancevalue of the variable resistor R12 has been adjusted to the small value.As a result, the negative gate current Ig(−) can be steeply varied inthe beginning stage of the transition time period T10 during which thetransistor 5 is turned OFF. Accordingly, the electron charges which havebeen stored in the gate electrode G of the transistor 5 can be quicklydischarged in the beginning stage of the transition time period T10 asto the turn-OFF of the transistor 5. As a result, such a time requiredfor turning OFF the transistor 5 in the beginning stage of thetransition time period T10 can be shortened. In some cases, as shown inFIG. 47A to FIG. 47E, the time for the transition time period T10required for turning OFF the transistor 5 can be shortened, as comparedwith the transition time period T100 of the conventional drivingcircuit.

When the operation is advanced to the final stage of the transition timeperiod T10 for turning OFF the transistor 5, the drain-to-source voltageVds is increased. The resistance value of the variable resistor R12 islargely adjusted in connection with the increase of the source-to-drainvoltage Vds. As a result, the negative gate current Ig(−) can be gentlyvaried in the final stage of the transition time period T10 during whichthe transistor 5 is turned OFF. Accordingly, the electron charges whichhave been stored in the gate electrode G of the transistor 5 can beslowly discharged in the final stage of the transition time period T10as to the turn-OFF of the transistor 5. As a result, the drain currentId of the transistor 5 gently flows, so that increasing of the surgevoltage can be suppressed.

In accordance with the driving circuit 410, in the transition timeperiod T10 during which the transistor 5 is turned OFF, a trade-offrelationship existed between the surge voltage and the turn-OFF loss canbe overcome.

Next, arrangements of concrete circuits will now be shown. It should benoted that the same reference numerals will be employed for denoting thesame, or similar structural elements, and therefore, explanationsthereof are omitted.

FIG. 48 indicates such an example that both the variable resistor R12and the drain voltage detecting means 450 shown in FIG. 46 are realizedby the pinch resistor 460. The pinch resistor 460 has been equipped withan n type first semiconductor region 462 of poly crystalline siliconcontaining an n type impurity, a p type semiconductor region 464 of polycrystalline silicon containing a p type impurity, and an n type secondsemiconductor region 465 of poly crystalline silicon containing an ntype impurity. The n type first semiconductor region 462 and the n typesecond semiconductor region 465 have been isolated from each other bythe p type semiconductor region 464. The n type first semiconductorregion 462 has been electrically connected via a first electrode 461 tothe drain electrode D of the transistor 5. The n type secondsemiconductor region 465 has been electrically connected via a secondelectrode 466 to the drain electrode D of the transistor 5. One end ofthe p type semiconductor region 464 has been electrically connected viaa third electrode 467 to the gate electrode G of the transistor 5. Theother end of the p type semiconductor region 464 has been electricallyconnected via a fourth electrode 463 to the gate terminal G10.

Referring now to FIG. 49A to FIG. 49C, such a technical idea that adepletion layer formed in the pinch resistor 460 is expanded/compressedwill be described. It should also be understood that as to impurityconcentration and a width of each of the semiconductor regions whichconstitute the pinch resistor 460, these semiconductor regions have beenformed in such a manner that the respective semiconductor regions aresubstantially completely depleted when a voltage corresponding to the DCvoltage Vps of the voltage supply source PS is applied between the firstelectrode 461 and the second electrode 466. Otherwise, as will bedescribed later, when a zener diode Dz has been provided between thepinch resistor 460 and the drain electrode D of the transistor 5, as toimpurity concentration and a width of each of the semiconductor regionswhich constitute the pinch resistor 460, these semiconductor regionshave been formed in such a manner that the respective semiconductorregions are substantially completely depleted so as to obtain adesirable resistance value when a voltage obtained by subtracting abreakdown voltage of the zener diode Dz from the DC voltage Vps of thevoltage supply source PS is applied between the first electrode 461 andthe second electrode 466.

In the case that the source-to-drain voltage Vds is 0 V, a depletionlayer has been formed based upon a diffusion potential differencebetween the p type semiconductor region 464 and the n type semiconductorregions 462 and 466. A width of this depletion layer is kept wide, and awidth of a current path of the p type semiconductor region 464 is keptwide.

In the case that the source-to-drain voltage Vds is 20 V, a pn junctionbetween the p type semiconductor region 464 and the n type semiconductorregions 462 and 466 is reverse-biased, so that the depletion layer isexpanded in the p type semiconductor region 464. As a result, the widthof the current path of the p type semiconductor region 464 becomesnarrows.

In the case that the source-to-drain voltage Vds is 75 V, the pnjunction between the p type semiconductor region 464 and the n typesemiconductor regions 462 and 466 is further reverse-biased, so that thedepletion layer is further expanded in the p type semiconductor region464. As a result, the width of the current path of the p typesemiconductor region 464 becomes narrows, and the p type semiconductorregion 464 becomes a high resistance.

FIG. 50 represents a relationship between a voltage V and a resistancevalue R of the pinch resistor 460, while the voltage V is appliedbetween the first electrode 461 and the second electrode 466 of thepinch resistor 460. As sown in FIG. 50, the resistance value R of thepinch resistor 460 is continuously increased in response to the voltageV to be applied.

As a result, the resistance value R of the pinch resistor 460 isadjusted to be small when the source-to-drain voltage Vds is low,whereas the resistance value R of the pinch resistor 460 is adjusted tobe large when the source-to-drain voltage Vds is high.

FIG. 51 shows a simulation result of the driving circuit 410 withemployment of the pinch resistor 460 having the above-describedcharacteristic. It should also be noted that the resistance value of thefixed resistor R10 was set to 3Ω. Also, in FIG. 52, a simulation resultobtained in such a case that the pinch resistor 460 is not provided isrepresented as a comparison example.

As shown in FIG. 51, in the driving circuit 410 with employment of thepinch resistor 460, it could be confirmed that the surge voltage isconspicuously suppressed.

FIG. 53 shows a trade-off curve existed between a surge voltage and aturn-OFF loss.

An indication “WITHOUT MEASURE” indicates a result of such a drivingcircuit that the pinch resistor 460 is not provided. Another indication“WITH MEASURE” shows a result of the driving circuit 410 where the pinchresistor 460 is employed.

In the case of “WITH MEASURE”, the surge voltage is conspicuouslyreduced without substantially increasing the turn-OFF loss, as comparedwith in the case of “WITHOUT MEASURE.” The result of “WITH MEASURE” canbe evaluated as follows: That is, a large improvement can be largelyachieved based upon the trade-off curve.

FIG. 54 shows an example as to a modification of the driving circuit410. This driving circuit 410 is featured by that a zener diode Dz hasbeen provided between the electrodes 461 and 466 of the pinch resistor460, and the drain electrode D of the transistor 5. An anode of thezener diode Dz has been connected to the side of the electrodes 461 and466 of the pinch resistor 460, and a cathode thereof has been connectedto the side of the drain electrode D of the transistor 5.

If the zener diode Dz is provided, then the source-to-drain voltage Vdsof the transistor 5 is not applied to the n type semiconductor regions462 and 465 of the pinch resistor 460 until the zener diode Dz breaksdown. Accordingly, until the zener diode Dz breaks down, the currentpath of the pinch resistor 460 is kept wide. As a result, in thebeginning stage of the transition time period during which thetransistor 5 is turned OFF, the resistance value of the pinch resistor460 can be kept small, and the negative gate current Ig(−) can besteeply varied. As a consequence, the drain current Id of the transistor5 can be steeply varied, so that a time required for turning OFF thetransistor 5 can be furthermore shortened.

Alternatively, a plurality of zener diodes Dz may be employed in aseries connection manner. The voltage applied between the n typesemiconductor regions 462 and 465 can be adjusted to a low voltage byconnecting the plurality of zener diodes Dz in the series connectionmanner. If the voltage applied between the n type semiconductor regions462 and 465 is adjusted to the low voltage, then the impurityconcentration of the p type semiconductor region 464 can be made low,and/or the width of the p type semiconductor region 464 interposedbetween the n type semiconductor regions 462 and 465 can be made narrow.Accordingly, the variation of the width of the depletion layer when thevoltage is applied between the n type semiconductor regions 462 and 465is increased, and thus, the resistance value of the p type semiconductorregion 464 is largely changed. As a result, the below-mentioned superioreffect can be achieved. That is, the resistance value of the pinchresistor 460 is kept small in the beginning stage of the transitionperiod during which the transistor 5 is turned OFF, whereas theresistance value of the pinch resistor 460 is kept large in the finalstage of the transition period during which the transistor 5 is turnedOFF.

Seventh Embodiment

FIG. 55 shows a circuit diagram of a driving circuit 410 ac cording to aseventh embodiment of the present disclosure. This driving circuit 410is featured by employing an MOS type resistor 560 having an MOSstructure.

The MOS type resistor 560 has been equipped with a first insulatorregion 562 made of a silicon oxide, a p type semiconductor region 564made of single crystalline silicon containing a p type impurity, and asecond insulator region 565 made of a silicon oxide. The first insulatorregion 562 and the second insulator region 565 have been isolated fromeach other by the p type semiconductor region 564. The first electrode561 has been located opposite to the p type semiconductor region 564 viathe first insulator region 562. The second electrode 566 has beenlocated opposite to the p type semiconductor region 564 via the secondinsulator region 565. Both the first electrode 561 and the secondelectrode 566 have been electrically connected to the drain electrode Dof the transistor 5. One end of the p type semiconductor region 564 hasbeen electrically connected via a third electrode 567 to the gateelectrode G of the transistor 5. The other end of the p typesemiconductor region 564 has been electrically connected via a fourthelectrode 563 to the gate terminal G10.

In the MOS type resistor 560, a width of a depletion layer which isexpanded/compressed within the p type semiconductor region 464 can beadjusted due to an electric field effect.

FIG. 56 represents a relationship between a voltage V and a resistancevalue R of the MOS type resistor 560, while the voltage V is appliedbetween the first electrode 561 and the second electrode 566 of the MOStype resistor 560. As shown in FIG. 56, the resistance value R of theMOS type resistor 560 is substantially continuously increased inresponse to the voltage V to be applied.

As a result, the resistance value R of the MOS type resistor 560 isadjusted to be small when the source-to-drain voltage Vds is low,whereas the resistance value R of the MOS type resistor 560 is adjustedto be large when the source-to-drain voltage Vds is high.

FIG. 57 shows a simulation result of the driving circuit 410 withemployment of the MOS type resistor 560 having the above-describedcharacteristic. It should also be noted that the resistance value of thefixed resistor R10 was set to 3Ω. Also, in FIG. 52, a simulation resultobtained in such a case that the MOS type resistor 560 is not providedis represented as a comparison example.

As shown in FIG. 57, in the driving circuit 410 with employment of theMOS type resistor 560, it could be confirmed that the surge voltage isconspicuously suppressed.

FIG. 58 shows a trade-off curve existed between a surge voltage and aturn-OFF loss.

An indication “WITHOUT MEASURE” indicates a result of such a drivingcircuit that the MOS type resistor 560 is not provided. Anotherindication “WITH MEASURE” shows a result of the driving circuit 410where the MOS type resistor 560 is employed.

In the case of “WITH MEASURE”, the surge voltage is conspicuouslyreduced without substantially increasing the turn-OFF loss, as comparedwith in the case of “WITHOUT MEASURE.” The result of “WITH MEASURE” canbe evaluated as follows: That is, a large improvement in the surgevoltage reduction can be largely achieved from the trade-off curve.

Eighth Embodiment

FIG. 59 to FIG. 62 indicate such an example that the following circuitelements of the driving circuit 410 of FIG. 54, namely, the pinchresistor 460, the zener diode Dz, the diode D10, the fixed resistor R10,and the protection-purpose zener diode 422 have been provided in thesame semiconductor substrate 20 is that of the transistor 5, accordingto an eighth embodiment of the present disclosure. Any of these circuitelements have been formed on the semiconductor substrate 20. Thetransistor 5 and these circuit elements have been formed in an integralform by utilizing the semiconductor substrate 20, and have been arrangedby a single chip.

It should be understood that in this eighth embodiment, the second diodeD12 within the driving circuit 410 shown in FIG. 54 has been deleted. Ifthe second diode D12 has been deleted, then such an effect can beachieved. That is, the positive gate current Ig(+) is gently varied inthe beginning stage of the turn-ON operation, whereas the positive gatecurrent Ig(+) is steeply varied in the final stage of the turn-ONoperation.

FIG. 59 is a perspective view for indicating a major portion of thepinch resistor 460 formed on the semiconductor substrate 20. It shouldalso be noted that reference numerals shown in FIG. 59 correspond toreference numerals applied to the respective structural elements of thepinch resistor 460 shown in FIG. 54. As will be described later, thepinch resistor 460 has been provided via an oxide film on thesemiconductor substrate 20. As a material of the pinch resistor 460,poly crystalline silicon has been employed. Phosphorus has beenconducted to both the n type first semiconductor region 462 and the ntype second semiconductor region 465, whereas boron has been conductedto the p type semiconductor region 464.

FIG. 60 indicates a surface layout of a semiconductor substrate 20 inwhich the transistor 5 and the circuit elements such as the pinchresistor 460 have been formed in an integral body. A portion indicatedby a broken line implies such a portion that an aluminum wiring line hasbeen formed. Referring now to the circuit diagram shown in FIG. 54, thesurface layout of FIG. 60 will be described.

A source electrode S of the transistor 5 has been arranged on a majorportion of a front surface of the semiconductor substrate 20.Semiconductor regions which are required for the transistor 5 have beenformed in the semiconductor substrate 20 under the source electrode S. Agate electrode G under such a condition that this gate electrode G iselectrically insulated from the source electrode S has been elongatedfrom a circumference portion of the source electrode S toward an insidethereof. The source electrode S and the gate electrode G of thetransistor 5 have been connected to each other via theprotection-purpose zener diode 422 at an upper left corner portion. Asthe material of the protection-purpose zener diode 422, poly crystallinesilicon has been employed. The protection-purpose zener diode 422 hasbeen provided via an oxide film on the semiconductor substrate 20.

The zener diode Dz, the pinch resistor 460, and the diode D10, and also,the fixed resistor R10 have been arranged at an upper right cornerportion of the surface layout. FIG. 61 is a longitudinal sectional viewcorresponding to a line “LXI-LXI” of FIG. 60. FIG. 62 is a longitudinalsectional view corresponding to a line “LXII-LXII” of FIG. 60.

As indicated in FIG. 61, a pinch-off resistor 460 has been provided viaan oxide film 472 on the semiconductor substrate 20. While an aluminumwiring line is extended above the pinch-off resistor 460, this aluminumwiring line has been electrically connected to both an n type firstsemiconductor region 462 and an n type second semiconductor region 465of the pinch-off resistor 460. This aluminum wiring line has beenelectrically insulated from the p type semiconductor region 464 of thepinch-off resistor 460. A portion to which the aluminum wiring line andthe n type first semiconductor region 462 are connected is referred toas a first electrode 461 which corresponds to the first electrode 461shown in FIG. 54. A portion to which the aluminum wiring line and the ntype second semiconductor region 465 are connected is referred to as asecond electrode 466 which corresponds to the second electrode 466 shownin FIG. 54. As shown in FIG. 60, one end of the aluminum line has beenelectrically connected to the cathode of the zener diode Dz. The anodeof the zener diode Dz has been connected to an equi-potential ringelectrode (EQR) which is provided around the semiconductor substrate 20in a circular manner. The equi-potential ring electrode (EQR) has beenelectrically connected to the drain electrode D of the transistor 5. Asa consequence, the anode of the zener diode Dz has been electricallyconnected via the equi-potential ring electrode (EQR) to the drainelectrode D of the transistor 5.

As indicated in FIG. 62, the p type semiconductor region 464 of thepinch-off resistor 460 has been electrically connected to two sets ofaluminum wiring lines at both terminals thereof. As represented in FIG.60, one aluminum wiring line has been extended from the gate terminalG10. This aluminum wiring line has been electrically connected via athird electrode 63 to one end of the p type semiconductor region 464. Asrepresented in FIG. 60, the other aluminum wiring line has beenelongated from the gate electrode G of the transistor 5. This aluminumwiring line has been electrically connected via a fourth electrode 67 tothe other end of the p type semiconductor region 464. The thirdelectrode 63 and the fourth electrode 67 correspond to the thirdelectrode 63 and the fourth electrode 67 of FIG. 54.

As shown in FIG. 60, a series circuit constructed of the diode D10 andthe fixed resistor R10 has been connected between the gate electrode Gand the gate terminal G10. As a material for the diode D10 and the fixedresistor R10, poly crystalline silicon has been employed. The resistancevalue of the fixed resistor R10 is adjusted based upon concentration ofa conducted impurity.

Since the above-described arrangement is employed, such circuit elementsas the pinch resistor 460, the zener diode Dz, the diode D10, the fixedresistor R10, and the protection-purpose zener diode 422 can bemanufactured in one integral body on the same semiconductor substrate asthat of the transistor 5. Since such an arrangement is employed, theabove-described respective circuit elements need not be separatelyprepared. As a consequence, the switching circuit-purpose chip disclosedin the specification of the present disclosure can be constructedwithout increasing a total number of these components. If the transistorand the above-described circuit elements are provided on the samesemiconductor substrate, then the switching circuit can be arranged byemploying a small number of the components. Thus, the switching circuitcan be furthermore made compact, and can be manufactured with superiorpractical characteristics.

Ninth Embodiment

FIG. 63 and FIG. 64 indicate an example of zener diodes Dz which havebeen connected between the drain electrode D of the transistor 5 and thepinch resistor 460, according to a ninth embodiment of the presentdisclosure.

In the example shown in FIG. 63, a plurality of zener diodes Dz havebeen connected between an n type first semiconductor region 462 of thepinch resistor 460 and the drain electrode D of the transistor 5.Furthermore, a plurality of zener diodes Dz have also been connectedbetween an n type second semiconductor region 465 of the pinch resistor460 and the drain D of the transistor 5. In other words, a parallelcircuit between the plurality of zener diodes Dz and the plurality ofzener diodes Dz has been connected between the drain electrode D of thetransistor 5 and the pinch resistor 460. In this example, a total numberof the above-described zener diodes Dz connected between the n typefirst semiconductor region 462 and the drain electrode D is equal tothat of the zener diodes Dz connected between the n type secondsemiconductor region 465 and the drain electrode D. Alternatively, asshown in FIG. 64, a series circuit constructed of the zener diodes Dzmay be combined between the drain electrode D and any one of the n typefirst conductor region 462 and the n type second conductor region 465.

If a plurality of zener diodes Dz are provided, then a voltage appliedbetween the n type semiconductor regions 462 and 465 can be adjusted toa lower voltage. In other words, the voltage applied between the n typesemiconductor regions 462 and 465 is adjusted to be such a voltage whichis obtained by subtracting a total value of breakdown voltages as to theplural zener diodes Dz from the source-to-drain voltage Vds of thetransistor 5. As a consequence, the voltage applied between the n typesemiconductor regions 462 and 465 can be adjusted in response to a totalnumber of zener diodes Dz. If the voltage applied between the n typesemiconductor regions 462 and 465 is adjusted to the low voltage, thenthe impurity concentration of the p type semiconductor region 464 can bemade low, and/or the width of the p type semiconductor region 464interposed between the n type semiconductor regions 462 and 465 can bemade narrow. Accordingly, the variation of the width of the depletionlayer when the voltage is applied between the n type semiconductorregions 462 and 465 is increased, and thus, the resistance value of thep type semiconductor region 464 is largely changed. As a result, thebelow-mentioned superior effect can be achieved. That is, the resistancevalue of the pinch resistor 460 is kept small in the beginning stage ofthe transition period during which the transistor 5 is turned OFF,whereas the resistance value of the pinch resistor 460 is kept large inthe final stage of the transition period during which the transistor 5is turned OFF.

Tenth Embodiment

A description is made of preferred features of a driving circuitaccording to a tenth embodiment of the present disclosure. The drivingcircuit of this tenth embodiment drives a field-effect type transistor.An adjusting-purpose transistor is a p type MOSFET. In this case, aresistor whose resistance value is larger than an ON resistance of the ptype MOSFET has been provided between a source electrode and a gateelectrode of the p type MOSFET. Furthermore, the source electrode of thep type MOSFET has been connected to the gate electrode of thetransistor; a drain electrode of the p type MOSFET has been grounded;and the gate electrode of the p type MOSFET has been connected to adriving voltage generating circuit. An adjusting-purpose transistor isan n type MOSFET. In this case, a resistor whose resistance value islarger than an ON resistance of the n type MOSFET has been providedbetween a source electrode and a gate electrode of the n type MOSFET.Furthermore, the source electrode and the gate electrode of the n typeMOSFET have been connected to the gate electrode of the transistor; andthe source electrode of the n type MOSFET has been connected to adriving voltage generating circuit.

FIG. 66 is a circuit diagram for showing a driving circuit 610 which isemployed so as to drive a field-effect type transistor 5. The FET typetransistor 5 has been connected between a load R1 and the ground (GND).A stray inductance of a wiring line and the like has been connectedbetween the transistor 5 and the load R1. The driving circuit 610applies a driving voltage “Vin” having a rectangular waveform to a gateelectrode G of the transistor 5, and switches turning-ON/OFF operationsof the transistor 5 based upon this driving voltage “Vin.” The drivingcircuit 610 switches the turning-ON/OFF operations of the transistor 5so as to switch such conditions that a DC voltage Vps of a voltagesupply source PS is applied, and is not applied to the load R1.

The driving circuit 610 has been equipped with a driving voltagegenerating circuit 411, a third resistor Rg3 (one example of fixedresistor), a first diode D10, and an adjusting circuit 620. The drivingvoltage generating circuit 411 has been electrically connected via thethird resistor Rg3 and the first diode D10 to the gate electrode G ofthe transistor 5. An anode of the first diode D10 has been connected tothe side of the driving voltage generating circuit 411, and a cathodethereof has been connected to the side of the gate electrode G of thetransistor 5. Both the first diode D10 and the adjusting circuit 620have constituted a parallel circuit between the driving voltagegenerating circuit 411 and the transistor 5. The adjusting circuit 620has been provided with a gate current adjusting circuit 622, a switchmeans SW, a first resistor Rg1, a second resistor Rg2, and a seconddiode D12. A resistance value of the first resistor Rg1 is smaller thana resistance value of the second resistor Rg2. An anode of the seconddiode D12 has been connected to the side of the gate electrode G of thetransistor 5, and a cathode thereof has been connected to the side ofthe driving voltage generating circuit 411.

The gate current adjusting circuit 622 detects a negative gate current“Ig(−)” when electron charges stored in the gate electrode G of thetransistor 5 are discharged, and switches open/close operations of theswitch means SW based upon a current value of the detected negative gatecurrent Ig(−). When an absolute value of the current value of thenegative gate current Ig(−) is large, the gate current adjusting circuit622 closes the switch means SW. When an absolute value of the currentvalue of the negative gate current Ig(−) is small, the gate currentadjusting circuit 622 opens the switch means SW.

FIG. 67A to FIG. 67E show operating waveform diagrams of the transistor5.

Firstly, a description is made of a transition time period during whichthe transistor 5 is turned ON. When a signal level of the drivingvoltage Vin is changed from a low level to a high level, the drivingvoltage Vin is converted into a positive gate current “Ig(+)” by thethird register Rg3. The positive gate current “Ig(+)” is supplied viathe first diode D10 to the gate electrode G of the transistor 5, sincethe second diode D12 has been provided in a reverse direction. When thepositive gate current “Ig(+)” is supplied to the gate electrode G of thetransistor 5, electron charges are stored in the gate electrode G. Whenthe electron charges are stored in the gate electrode G, agate-to-source voltage “Vgs” of the transistor 5 is increased. When thegate-to-source voltage “Vgs” of the transistor 5 is increased, a draincurrent “Id” starts to flow from the drain electrode D of the transistor5 toward the source electrode S, so that the drain-to-source voltage Vdsis decreased. The state of the transistor 5 is transferred from the OFFstate to the ON state by performing these operation steps.

Next, a description is made of a transition time period “T10” duringwhich the transistor 5 is turned ON. When the signal level of thedriving voltage Vin becomes a low level from the high level, theelectron charges which have been stored in the gate electrode G aredischarged. Since the first diode D10 has been provided, a negative gatecurrent “Ig(−)” produced in connection with the discharge of theelectron charges flows toward the gate current adjusting circuit 622. Anabsolute value of the current value of the negative gate current “Ig(−)”is gradually increased in a beginning stage T12 of the transition timeperiod T10 during which the transistor 5 is turned OFF. The gate currentadjusting circuit 622 detects the absolute value of the current value ofthe negative gate current “Ig(−)”, and if this detected absolute valueexceeds a predetermined threshold value (namely, if current valuebecomes lower than threshold value), then the gate current adjustingcircuit 622 closes the switch means SW. As a consequence, the negativegate current Ig(−) flows through the first resistor Rg1 having the smallresistance value to the ground GND. When the switch means SW is closed,in the beginning stage T12 of the transition time period T10 duringwhich the transistor 5 is turned OFF, the negative gate current “Ig(−)”may behave in such a manner that the absolute value of this currentvalue is rapidly increased. As a consequence, the adjusting circuit 620is operated in such a manner that the electron charges are quicklydischarged from the gate electrode G of the transistor 5 in thebeginning stage T12 of the transition time period T10 during which thetransistor 5 is turned OFF. As a result, such a time required forturning OFF the transistor 5 in the beginning stage T12 of thetransition time period T10 can be shortened. In some cases, as shown inFIG. 67A to FIG. 67E, the time for the transition time period T10required to turn OFF the transistor 5 can be shortened, as compared withthe transition time period T100 of the conventional driving circuit.

It should also be understood that until the absolute value of thecurrent value of the negative gate current Ig(−) exceeds a predeterminedthreshold value, the switch means SW has been opened, so that theresistance value of the gate resistor is large and the electron chargesare slowly discharged. However, this time period is extremely short, andtherefore, does not essentially prolong the time required for turningOFF the transistor 5.

When the operation is advanced to a final stage T14 of the transitiontime period T10 for turning OFF the transistor 5, the current value ofthe negative gate current Ig(−) is gradually decreased. The gate currentadjusting circuit 622 detects the absolute value of the current value ofthe negative gate current “Ig(−)”, and if this detected absolute valuebecomes lower than a predetermined threshold value (namely, if currentvalue becomes higher than threshold value), then the gate currentadjusting circuit 622 opens the switch means SW. As a consequence, thenegative gate current Ig(−) flows through the second resistor Rg2 havingthe large resistance value, the third resistor Rg3, and the drivingvoltage generating circuit 411 to the ground GND. When the switch meansSW is opened, in the final stage T14 of the transition time period T10during which the transistor 5 is turned OFF, the negative gate current“Ig(−)” may behave in such a manner that the absolute value of thiscurrent value is decreased. As a consequence, in the final stage T14 ofthe transition time period T10 of turning off, the adjusting circuit 620is operated in such a manner that the electron charges are slowlydischarged from the gate electrode G of the transistor 5. As a result,the drain current Id of the transistor 5 gently flows, so thatincreasing of the surge voltage can be suppressed.

In accordance with the driving circuit 610, in the transition timeperiod T10 during which the transistor 5 is turned OFF, a trade-offrelationship existed between the surge voltage and the turn-OFF loss canbe overcome.

Also, the driving circuit 610 of this tenth embodiment has such afeature that the gate current Ig is monitored. Generally speaking, thereare many possibilities that a plurality of loads have been connectedbetween the voltage supply source PS and the transistor 5. As a result,if a large current flows through a certain load among these pluralloads, then the DC voltage Vps of the voltage supply source PS islowered. More specifically, in the automobile field, since batteries donot contain sufficiently large current capacities, the DC voltage Vps ofthe voltage supply source PS may be easily varied.

For example, the following technical solution may be conceived. That is,a drain voltage of the transistor 5 is monitored so as to adjust thegate resistor thereof. However, in the case where the drain voltage ofthe transistor 5 is monitored, the DC voltage Vps of the voltage supplysource PS is varied, so that the drain voltage of the transistor 5 issimilarly varied. As a result, final stage timing of a transition timeperiod during which a transistor is turned OFF cannot be correctlygrasped.

On the other hand, in such a case where the gate current Ig is monitoredby the driving circuit 610 of the tenth embodiment, even when the DCvoltage Vps of the voltage supply source PS is varied, the gate currentIg is not adversely influenced by this voltage variation. As a result,the final stage timing of the transition time period during which thetransistor is turned OFF can be correctly grasped.

FIG. 68 shows an example in which the gate current adjusting circuit622, the switch means SW, and the first resistor Rg1 shown in FIG. 66have been realized by a p type MOSFET (Metal Oxide SemiconductorField-Effect Transistor) 624. Both a threshold voltage and a switchingoperation of the p type MOSFET 624 are equivalent to the gate currentadjusting circuit 622 and the switch means SW shown in FIG. 66. An ONresistance of the p type MOSFET 624 is equivalent to the first resistorRg1.

The second resistor Rg2 have been provided between a gate electrode Gand a source electrode S (one example of input electrode) of the p typeMOSFET 624. As a consequence, the p type MOSFET 624 is turned ON/OFF inresponse to a voltage difference produced between both terminals of thesecond resistor Rg2. Further, one terminal of the second resistor Rg2has been connected to the gate electrode G of the transistor 5, and theother terminal thereof has been connected via the third resistor Rg3,and the driving voltage generating circuit 411 to the ground GND. Thedrain electrode D of the p type MOSFET 624 has been connected to theground GND.

In accordance with the above-described driving circuit 610, within thetransition time period during which the transistor 5 is turned OFF, inthe beginning stage where the absolute value of the current value of thenegative gate current Ig(−) becomes large, the p type MOSFET 624 isturned ON in response to the voltage difference produced between boththe terminals of the second resistor Rg2. When the p type MOSFET 624 isturned ON, the negative gate current Ig(−) passes from the sourceelectrode S of the p type MOSFET 624 through the drain electrode D tothe ground GND. Since the ON resistance of the p type MOSFET 624 issmall, the electron charges which have been stored in the gate electrodeG of the transistor 5 can be quickly discharged. As a result, a timerequired for turning OFF the transistor 5 can be shortened.

On the other hand, in the final stage where the absolute value of thecurrent value of the negative gate current Ig(−) becomes small, thevoltage difference produced between both the terminals of the secondresistor Rg2 becomes low, so that the p type MOSFET 624 is turned OFF.When the p type MOSFET 624 is turned OFF, the negative gate currentIg(−) flows through the second resistor Rg2, the third resistor Rg3, andthe driving voltage generating circuit 411 to the ground GND. Since theresistance value of the second resistor Rg2 is large, the electroncharges which have been stored in the gate electrode G of the transistor5 can be slowly discharged. As a result, increasing of the surge voltagecan be suppressed.

FIG. 69 represents a variation of a source-to-drain voltage Vds in thetransition time period during which the transistor 5 is turned OFF. Anindication “WITH P-MOS” indicates a result obtained of the drivingcircuit 610 shown in FIG. 68. An indication “WITHOUT MEASURE” indicatesa result obtained when only the first resistor Rg1 of FIG. 66 isprovided. An indication “WITHOUT P-MOS” shows a result when only thesecond resistor Rg2 of FIG. 68 is provided. It should also be noted thatthe resistance value of the first resistor Rg1 is 0.1Ω; the resistancevalue of the second resistor Rg2 is 30Ω; and the resistance value of thethird resistor Rg3 is 3Ω. Also, a threshold voltage Vth of the p typeMOSFET 624 is −2.7 V.

As shown in FIG. 69, in the case of “WITHOUT MEASURE”, a speed fordischarging the electron charges is high, and a high surge voltage isgenerated. On the other hand, in case of “WITH P-MOS” and “WITHOUTP-MOS”, the speed for discharging the electron charges is suppressed tolow speed, and increasing of the surge voltage is suppressed. However,in case of “WITHOUT P-MOS”, the time required to turn OFF the transistoris prolonged, and the turn-OFF loss is increased.

FIG. 70 shows a relationship between a surge voltage and a turn-OFFloss.

In case of “WITHOUT P-MOS”, although the surge voltage is suppressed, ascompared with “WITHOUT MEASURE”, the time required for turning OFF thetransistor 5 becomes long, and the turn-OFF loss is increased. It is notpossible to evaluate from the trade-off curve that a result of “WITHOUTP-MOS” is largely improved.

On the other hand, in case of “WITH P-MOS”, while the turn-OFF loss isnot substantially increased, the surge voltage is conspicuously reduced,as compared with such a case of “WITHOUT MEASURE.” It is possible toevaluate from the trade-off curve that a result of “WITH P-MOS” islargely improved.

FIG. 71 shows a variation of the source-to-drain voltage Vds when theresistance value of the second resistor Rg2 is changed to 3Ω, 10Ω, and30Ω.

The larger the resistance value of the second resistor Rg2 is increased,the larger the speed for discharging the electron charges in the finalstage of the transition time period is suppressed, so that increasing ofthe surge voltage can be suppressed.

FIG. 72 shows a relationship between a surge voltage and a turn-OFF losswhen the resistance value of the second resistor Rg2 is changed to 3Ω,10Ω, and 30 Ω.

The larger the resistance value of the second resistor Rg2 is increased,the larger the trade-off relationship between the turn-OFF loss and thesurge voltage can be overcome.

FIG. 73 indicates a variation of the source-to-drain voltage Vds whenthe threshold voltage Vth of the p type MOSFET 624 is changed to −1.7 V,−2.7 V, and −3.7 V.

The smaller the absolute value of the threshold voltage Vth of the ptype MOSFET 624 is decreased, the longer discharging of the electroncharges via the p type MOSFET 624 is maintained. As a result, the timerequired for turning OFF the transistor 5 can be shortened. Furthermore,the electron charges are discharged via the second resistor Rg2 in thefinal stage of the transition time period for turning OFF the transistor5, so that increasing of the surge voltage can be suppressed.

FIG. 74 indicates a relationship between a surge voltage and a turn-OFFloss when the threshold voltage Vth of the p type MOSFET 624 is changedto −1.7 V, −2.7 V, and −3.7 V.

In the case that the threshold voltage Vth of the p type MOSFET 624 isadjusted, even if any of these threshold voltages Vth is employed, thenthe trade-off relationship existed between the surge voltage and theturn-OFF loss can be overcome.

FIG. 75 shows an example in which the gate current adjusting circuit622, the switch means SW, and the first resistor Rg1 shown in FIG. 66have been realized by an n type MOSFET 626. Both a threshold voltage anda switching operation of the n type MOSFET 626 are equivalent to thegate current adjusting circuit 622 and the switch means SW shown in FIG.66. An ON resistance of the n type MOSFET 626 is equivalent to the firstresistor Rg1.

The second resistor Rg2 have been provided between a gate electrode Gand a source electrode S (one example of input electrode) of the n typeMOSFET 626. As a consequence, the n type MOSFET 626 is turned ON/OFF inresponse to a voltage difference produced between both terminals of thesecond resistor Rg2. Further, one terminal of the second resistor Rg2has been connected to the gate electrode G of the transistor 5, and theother terminal thereof has been connected via the second diode D12, thethird resistor Rg3, and the driving voltage generating circuit 411 tothe ground GND. The drain electrode D and the gate electrode G of the ntype MOSFET 626 has been connected between one terminal of the secondresistor Rg2 and the gate electrode G of the transistor 5. The seconddiode D12 has been provided between the driving voltage generatingcircuit 411 and the source electrode S of the n-MOSFET 626.

In accordance with the above-described driving circuit 610, within thetransition time period during which the transistor 5 is turned OFF, inthe beginning stage where the absolute value of the current value of thenegative gate current Ig(−) becomes large, the n type MOSFET 626 isturned ON in response to the voltage difference produced between boththe terminals of the second resistor Rg2. When the n type MOSFET 626 isturned ON, the negative gate current Ig(−) passes from the drainelectrode D of the n type MOSFET 626 through the source electrode S, andfurthermore, flows through the second diode D12, the third resistor Rg3and the driving voltage generating circuit 411 to the ground GND. Inother words, the negative gate current Ig(−) flows by bypassing thesecond resistor Rg2. Since the ON resistance of the n type MOSFET 626 issmall, the electron charges which have been stored in the gate electrodeG of the transistor 5 can be quickly discharged. As a result, a timerequired for turning OFF the transistor 5 can be shortened.

On the other hand, in the beginning stage where the absolute value ofthe current value of the negative gate current Ig(−) becomes small, thevoltage difference produced between both the terminals of the secondresistor Rg2 becomes low, so that the n type MOSFET 626 is turned OFF.When the n type MOSFET 625 is turned OFF, the negative gate currentIg(−) flows through the second resistor Rg2, and further flows throughthe second diode D12, the third resistor Rg3, and the driving voltagegenerating circuit 411 to the ground GND. Since the resistance value ofthe second resistor Rg2 is large, the electron charges which have beenstored in the gate electrode G of the transistor 5 can be slowlydischarged. As a result, increasing of the surge voltage can besuppressed.

FIG. 76 shows a variation of the source-to-drain voltage Vds when theresistance value of the second resistor Rg2 is changed to 3Ω, 10Ω, and30Ω. In this case, a threshold voltage Vth of the n type MOSFET 626 is1.1 V.

The larger the resistance value of the second resistor Rg2 is increased,the larger the speed for discharging the electron charges in the finalstage of the transition time period is suppressed, so that increasing ofthe surge voltage can be suppressed.

FIG. 77 shows a relationship between a surge voltage and a turn-OFF losswhen the resistance value of the second resistor Rg2 is changed to 3Ω,10Ω, and 30 Ω.

The larger the resistance value of the second resistor Rg2 is increased,the trade-off relationship existed between the surge voltage and theturn-OFF loss can be overcome.

FIG. 78 shows a variation of the source-to-drain voltage Vds when thethreshold voltage Vth of the n type MOSFET 626 is changed to 0.1 V, 1.1V, and 2.1 V.

The lower the threshold voltage Vth of the n type MOSFET 626 becomes,discharging of the electron charges via the n type MOSFET 626 ismaintained for a long time, so that the time required for turning OFFthe transistor 5 is shortened. Moreover, in the final stage of thetransition time period for turning OFF the transistor 5, the electroncharges are discharged via the second resistor Rg2. As a result,increasing of the surge voltage can be suppressed.

FIG. 79 shows a relationship between a surge voltage and a turn-OFF losswhen the threshold voltage Vth of the n type MOSFET 626 is changed to0.1 V, 1.1 V, and 2.1 V.

In such a case that the threshold voltage Vth of the n type MOSFET 626is adjusted, even if any of these threshold voltages Vth is employed,then the trade-off relationship existed between the surge voltage andthe turn-OFF loss can be overcome.

The above-disclosed inventive ideas of the present disclosure cover thebelow-mentioned various modification modes.

In accordance with a first modification mode of the present disclosedideas, a switching circuit is arranged by a transistor having a firstelectrode, a second electrode, and a control electrode; a zener diode;and a capacitor. Since a switching voltage of the transistor isswitched, a junction between the first electrode and the secondelectrode can be temporally switched between a conduction state and anon-conduction state. The zener diode has been series-connected to thecapacitor between the first electrode and the control electrode of thetransistor. The first electrode is either a drain or a collector.

Concretely speaking, a cathode side of the zener diode is connected tothe drain (or collector), and an anode side of the zener diode isconnected to the control electrode. In this case, the capacitor may bealternatively inserted between the zener diode and the drain (orcollector), or the capacitor may be alternatively inserted between thezener diode and the control electrode.

In the above-described switching circuit, the zener diode has beeninserted between the drain (or collector) and the control electrode. Asa consequence, just after the transistor is turned OFF, while a voltagebetween the drain (or collector) and the control electrode is lower thana zener voltage (namely, breakdown voltage) of the zener diode, thiscondition is nearly equal to such a condition that the series circuitconstructed of the zener diode and the capacitor is not present. As aresult, a capacitance between the drain (or collector) and the controlelectrode is maintained under small capacitance condition; and both acurrent flowing between the drain and the source (or collector andemitter) of the transistor, and a voltage between them are rapidlychanged, so that a switching loss can be suppressed to a small loss.

A voltage between the drain and the source (or collector and emitter) isincreased after the transistor has been turned OFF, and in conjunctionwith this voltage increase, a voltage between the drain (or collector)and the control electrode is also increased. When the voltage betweenthe drain (or collector) and the control electrode exceeds the zenervoltage of the zener diode, the drain (or collector) is connected viathe capacitor to the control electrode, so that a capacitance between bedrain (or collector) and the control electrode is increased. As aconsequence, thereafter, both a current flowing between the drain andthe source (or collector and emitter) and a voltage between them areslowly changed. As a consequence, a surge voltage can be suppressed to alow voltage.

Accordingly, in accordance with the above-described switching circuit,both the switching loss and the surge voltage can be suppressed to thelower values.

Also, an internal structure of the transistor need not be improved, orthe internal structure thereof may be suppressed at such a degree thatthe internal structure is improved (if necessary, manner capable ofimproving internal structure of transistor may be alternatively combinedwith above-described switching circuit). As a result, a transistorhaving a low ON resistance (ON voltage) may be employed. Also, since theabove-described switching circuit may be realized by merely adding thezener diode and the capacitor, this switching circuit may be arranged byemploying a small number of circuit elements. Both the zener diode andthe capacitor may be readily manufactured at the same time in asemiconductor substrate used to form a transistor when the transistor ismanufactured. If the transistor, the zener di ode, and the capacitor areconstituted in an internal body in the same semiconductor substrate,then a total number of the circuit elements of the switching circuit isnot increased.

Alternatively, as an alternative idea, the above-described transistormay be realized by either a MOSFET or an IGBT. In this case, thetransistor may be realized as a unipolar transistor, or a bipolartransistor.

Alternatively, as an alternative idea, when the transistor is under OFFstate, a predetermined voltage is applied between the first electrodeand a second electrode, and the zener diode may have a zener voltagehigher than the predetermine voltage by approximately 0.5 to 1.0 times.In this case, the expression “OFF state” implies such a time period thata current change and a voltage change occurred after the transistorturned OFF have been converged. In the case that the above-describedcondition can be satisfied, both the switching loss and the surgevoltage can be suppressed under better balancing condition.

As an alternative idea, a transistor, a zener diode, and a capacitor maybe alternatively arranged on the same semiconductor substrate. In thisalternative case, the expression “circuit elements have been provided inthe same semiconductor substrate” involves such a case that when therespective circuit elements have been formed within the semiconductorsubstrate, either a portion or all of these circuit elements have beenmanufactured on the semiconductor substrate. In other words, theexpression “circuit elements have been provided in same semiconductorsubstrate” implies such a condition that the circuit elements as to thetransistor, the zener diode, and the capacitor have been constructed inone integral body by utilizing the semiconductor substrate. In thiscase, a total number of the circuit components which constitute theswitching circuit is not increased. Also, the transistor, the zenerdiode, and the capacitor may be easily formed in the semiconductorsubstrate by utilizing a semiconductor manufacturing technique. Thetransistor, the zener diode, and the capacitor may be formed in the samesemiconductor substrate without any forcible manner. Also, the switchingcircuit can be made compact and can be practically utilized under bettercondition. Furthermore, the zener diode may be alternatively providedvia an insulating film on the semiconductor substrate. The zener diodehas a p conductive type anode semiconductor region and an n conductivetype cathode semiconductor region, while the cathode semiconductorregion is contacted to the anode semiconductor region. It should beunderstood that both the anode semiconductor region and the cathodesemiconductor region may be made of poly crystalline silicon.

As an alternative idea, the capacitor may alternatively has an embeddedconductive region and a coating insulation region. The embeddedconductive region has been elongated from a first plane of asemiconductor substrate to a second plane thereof. The coatinginsulation region coats the embedded insulation region in such a mannerthat the embedded conductive region is electrically insulated from thesemiconductor substrate. In this case, the coating insulation region mayfunction as a dielectric material which constitutes the capacitor.

As an alternative idea, the zener diode may alternatively has a pconductive type anode semiconductor region and an n conductive typecathode semiconductor region. The anode semiconductor region has beenprovided via an insulating film on the semiconductor substrate. Thecathode semiconductor region has been provided via the insulating filmon the semiconductor substrate, and is contacted to the anodesemiconductor region. The capacitor has the embedded conductive regionand the coating insulation region. The embedded conductive region hasbeen elongated from the first plane to the second plane of thesemiconductor substrate. The coating insulation region has coated theembedded conductive region in such a manner that the embedded conductiveregion is electrically insulated from the semiconductor substrate. Theanode semiconductor region has been electrically connected to thecontrol electrode of the transistor. The cathode semiconductor regionhas been electrically connected to the embedded conductive region. Theembedded conductive region has been electrically insulated from thefirst electrode of the transistor by the coating insulation region.Furthermore, the capacitor may alternatively and further have adiffusion semiconductor region arranged within the semiconductorsubstrate. The coating insulation region has been sandwiched between theembedded conductive region and the diffusion semiconductor region. Thediffusion semiconductor region has been electrically connected to thefirst electrode of the transistor. The second electrode of thetransistor corresponds to either a source or an emitter.

As an alternative idea, the zener diode may alternatively has a pconductive type anode semiconductor region and an n conductive typecathode semiconductor region. The anode semiconductor region has beenprovided via an insulating film on the semiconductor substrate. Thecathode semiconductor region has been provided via the insulating filmon the semiconductor substrate, and is contacted to the anodesemiconductor region. The capacitor has the embedded conductive regionand the coating insulation region. The embedded conductive region hasbeen elongated from the first plane to the second plane of thesemiconductor substrate. The coating insulation region has coated theembedded conductive region in such a manner that the embedded conductiveregion is electrically insulated from the semiconductor substrate. Theembedded conductive region has been electrically connected to thecontrol electrode of the transistor. The anode semiconductor region hasbeen electrically insulated from the embedded conductive region by thecoating insulation region. The cathode semiconductor region has beenelectrically connected to the first electrode of the transistor.Moreover, the capacitor may alternatively and further have a diffusionsemi-conductor region arranged within the semiconductor substrate. Thecoating insulation region has been sandwiched between the embeddedconductive region and the diffusion semiconductor region. The diffusionsemiconductor region has been electrically connected to the anodeconductor region. The second electrode of the transistor corresponds toeither a source or an emitter.

In accordance with a second modification mode of the present disclosedideas, a switching circuit is arranged by a transistor having a firstelectrode, a second electrode, and a control electrode; a zener diode;and a capacitor. Since a switching voltage of the transistor isswitched, a junction between the first electrode and the secondelectrode can be temporally switched between a condition state and anon-conduction state. The zener diode has been series-connected to thecapacitor between the first electrode and the second electrode of thetransistor. The first electrode is either a drain or a collector. Thesecond electrode is either a source of an emitter.

Concretely speaking, a cathode of the zener diode is connected to thedrain (or collector), and an anode of the zener diode is connected tothe source (or emitter). In this case, the capacitor may bealternatively inserted between the zener diode and the drain (orcollector), or the capacitor may be alternatively inserted between thezener diode and the source (or emitter).

Also, in this case, while the zener diode does not break down, both acurrent flowing between the drain and the source (or collector andemitter) of the transistor, and a voltage between them are rapidlychanged, so that a switching loss can be suppressed to a small loss.Thereafter, since the zener diode breaks down, both a current flowingbetween the drain and the source (or collector and emitter) and avoltage between them are slowly changed. As a consequence, a surgevoltage can be suppressed to a low surge voltage.

Both the switching loss and the surge voltage can be suppressed to thelow values even by the switching circuit of the above-described case.Even by this switching circuit, the ON resistance can be suppressed tothe low value, the switching loss can be suppressed to the small value,and the surge voltage can be suppressed to the low voltage. Moreover,the switching circuit may be arranged by employing a small number ofcircuit elements. If the transistor, the zener diode, and the capacitorare constituted in an integral body in the same semiconductor substrate,then a total number of the circuit elements of the switching circuit isnot increased.

As an alternative idea, the transistor, the zener diode, and thecapacitor may be alternatively arranged in the same semiconductorsubstrate. In addition, the capacitor may alternatively have an embeddedconductive region and a coating insulation region. The embeddedconductive region has been elongated from a first plane of asemiconductor substrate to a second plane thereof. The coatinginsulation region coats the embedded insulation region in such a mannerthat the embedded conductive region is electrically insulated from thesemiconductor substrate. It should also be noted that the zener diodemay be alternatively formed via an insulating film on the semiconductorsubstrate. Also, the zener diode may alternatively have a p conductivetype anode semiconductor region and an n conductive type cathodesemiconductor region. The cathodes semiconductor region has beencontacted to the anode semiconductor region. The cathode semiconductorregion has been electrically connected to the first electrode of thetransistor. In addition, the capacitor has a diffusion semiconductorregion arranged within the semiconductor substrate. The coatinginsulation region has been sandwiched between the embedded conductiveregion and the diffusion semiconductor region. The diffusionsemiconductor region has been electrically connected to the anodesemiconductor region. The embedded conductive region has beenelectrically connected to the second electrode of the transistor.

In accordance with a third modification mode of the present disclosedideas, a switching circuit is arranged by a transistor, a controlcircuit, a series circuit, and a voltage adjusting circuit. Thetransistor has a high voltage-sided major electrode, a low voltage-sidedmajor electrode, and a control electrode. In the control circuit, both apower supply and a load are series-connected between the highvoltage-sided major electrode and the low voltage-sided major electrode,and the control circuit outputs a voltage for turning ON/OFF thetransistor. While the control circuit is connected to the controlelectrode of the transistor, the series circuit has a first capacitorand a first diode. While the series circuit is connected between thecontrol electrode of the transistor and the high voltage-sided majorelectrode, the first capacitor has been series-connected to the firstdiode; a cathode of the first diode has been connected to the controlelectrode side; and an anode of the first diode has been connected tothe high voltage-sided major electrode. Then, the voltage adjustingcircuit has been connected to a junction portion between the firstcapacitor and the first diode. The voltage adjusting circuit adjusts avoltage of the junction portion.

It should be noted that in the series circuit constructed of the firstcapacitor and the first diode, the first capacitor may be arranged onthe control electrode side of the transistor, and the first diode may bearranged on the major electrode side of the high voltage side of thetransistor. Alternatively, the above-described positional relationshipmay be reversed.

In the above-described switching circuit, when the transistor is turnedOFF, if a forward direction voltage starts to flow through the firstdiode, then a charging current starts to flow into the first capacitor.Thereafter, a changing speed of the voltage at the control electrode ofthe transistor is slowed; a changing speed of the main current of thetransistor is slowed; and a voltage changing speed of the majorelectrode of the high voltage side of the transistor is slowed, so thata surge voltage appeared at the major electrode of the high voltage sideof the transistor can be suppressed to the low surge voltage.

If the voltage adjusting circuit is not provided, then such a timingwhen the forward direction voltage starts to effect the first diode isfixed to another timing when the voltage at the major electrode of thehigh voltage side of the transistor is increased up to the power supplyvoltage.

To the contrary, if the voltage adjusting circuit is additionallyprovided, then the forward direction voltage can be commenced to effectthe first diode at such a timing which is earlier than the timing whenthe voltage at the major electrode of the high voltage side of thetransistor is increased up to the power supply voltage; the changingspeed of the voltage at the control electrode of the transistor can beslowed at the earlier timing; the changing speed of the main current ofthe transistor can be slowed at the earlier timing; and the voltagechanging speed of the major electrode of the high voltage side of thetransistor can be slowed at the earlier timing. As a result, the surgevoltage appeared on the major electrode of the high voltage side of thetransistor can be suppressed to the low surge voltage.

If required, the forward direction voltage may be commenced to effectthe first diode at such a timing when the voltage at the major electrodeof the high voltage side of the transistor is increased higher than, orequal to the power supply voltage; the changing speed of the voltage atthe control electrode of the transistor may be slowed at the slowertiming; the changing speed of the main current of the transistor may beslowed at the slower timing; and the voltage changing speed of the majorelectrode of the high voltage side of the transistor may be slowed atthe slower timing. In this case, the switching loss of the transistorcan be reduced.

As previously described, since the voltage adjusting circuit isutilized, such a timing can be adjusted which slows the changing speedof the voltage generated at the control electrode of the transistor whenthe transistor is turned OFF. If this timing is adjusted to becomeearlier by the voltage adjusting circuit, then the surge voltage can besuppressed to the low surge voltage.

In a beginning stage during which the transistor is turned OFF, achanging speed of a voltage generated at the control electrode of thetransistor is increased so as to suppress a turn-OFF loss (since voltagebetween major electrodes of transistor is low in beginning stage, thereis no opportunity that surge voltage becomes excessively high, so thatchanging speed of voltage at control electrode can be increased) whereasin a final stage (namely, time period during which voltage between majorelectrodes of transistor has been increased, and thus, surge voltage canbecome excessively high) during which the transistor is turned OFF, thechanging speed of the voltage generated at the control electrode of thetransistor is slowed, so that the surge voltage can be suppressed to thelow surge voltage.

Generally speaking, there is a trade-off relationship between a turn-OFFloss and a surge voltage. If the turn-OFF loss is suppressed, then thesurge voltage becomes excessively high, whereas if the surge voltage issuppressed, then the turn-OFF loss becomes excessively large. In theabove-described circuit, while the transistor is turned OFF, thechanging speed of the voltage generated at the control electrode of thetransistor is slowed. As a result, both the turn-OFF loss and the surgevoltage can be suppressed.

Alternatively, as an alternative idea, in such a case that thetransistor is turned ON, the first capacitor may have an ON-statecharging voltage. Alternatively, in such a case that the transistor isturned OFF, the first capacitor many have an OFF-state charging voltage,and the voltage adjusting circuit adjusts the ON-state charging voltageto become lower than the OFF-state charging voltage. In accordance withthe above-described modification mode, the forward direction voltage maybe commenced to effect the first diode at such a timing which is earlierthan the timing when the voltage at the major electrode of the highvoltage side of the transistor is increased up to the power supplyvoltage; the changing speed of the voltage at the control electrode ofthe transistor may be slowed at the earlier timing; the changing speedof the main current of the transistor may be slowed at the earliertiming; and the voltage changing speed of the major electrode of thehigh voltage side of the transistor may be slowed at the earlier. As aresult, the surge voltage appeared on the major electrode of the highvoltage side of the transistor may be suppressed to the low surgevoltage.

Alternatively, as an alternative idea, in the case that the voltage atthe high voltage-sided major electrode of the transistor is lowered, thefirst capacitor may have a first charging voltage. Also, in the casethat the voltage at the high voltage-sided major electrode of thetransistor is equal to the power supply voltage, the first capacitor mayhave a second charging voltage, and the voltage adjusting circuitadjusts the first charging voltage to become lower than the secondcharging voltage. Even in accordance with this modification mode, theforward direction voltage may be commenced to effect the first diode atsuch a timing which is earlier than the timing when the voltage at themajor electrode of the high voltage side of the transistor is increasedup to the power supply voltage; the changing speed of the voltage at thecontrol electrode of the transistor may be slowed at the earlier timing;the changing speed of the main current of the transistor may be slowedat the earlier timing; and the voltage changing speed of the majorelectrode of the high voltage side of the transistor may be slowed atthe earlier timing. As a result, the surge voltage appeared on the majorelectrode of the high voltage side of the transistor can be suppressedto the low surge voltage.

Alternatively, as an alternative idea, the voltage adjusting circuit maycontain a second capacitor and a second diode. Both the second capacitorand the second diode have been series-connected between the junctionportion of the first capacitor and the first diode, and the anode of thefirst diode. A cathode of the second diode has been connected to theanode side of the first diode. An anode of the second diode has beenconnected to the junction portion side. In other words, both the secondcapacitor and the second diode have provided a signal path for bypassingthe first diode. In this voltage adjusting circuit, when the transistoris advanced to the ON state and thus the voltage at the major electrodeof the high voltage side of the transistor is decreased, a forwarddirection voltage is effected to the second diode. As a result, aportion of the electron charges which have been charged to the firstcapacitor is moved in such a manner that the second capacitor ischarged, so that the voltage at the connection line for connecting thefirst capacitor to the first diode is lowered. The voltage at theconnection line becomes a high voltage when the transistor is turnedOFF, and becomes a low voltage when the transistor is turned ON. In thiscase, the forward direction voltage may be commenced to effect the firstdiode at such a timing which is earlier than the timing when the voltageat the major electrode of the high voltage side of the transistor isincreased up to the power supply voltage; the changing speed of thevoltage at the control electrode of the transistor may be slowed at theearlier timing; the changing speed of the main current of the transistormay be slowed at the earlier timing; and the voltage changing speed ofthe major electrode of the high voltage side of the transistor may beslowed at the earlier timing. As a result, the surge voltage appeared onthe major electrode of the high voltage side of the transistor may besuppressed to the low surge voltage.

In addition, the second capacitor may alternatively have anelectrostatic capacity smaller than, or equal to the electrostaticcapacity of the first capacitor. If the electrostatic capacity of thesecond capacitor becomes excessively large, then an amount of electroncharges stored in the first capacitor becomes excessively small, so thata charging voltage of the first capacitor becomes excessively low. Ifthe charging voltage of the first capacitor becomes excessively low,then a forward direction voltage starts to effect the first diode fromthe beginning stage of the transition time period during which thetransistor is turned OFF, so that a turn-OFF loss is increased. If theelectrostatic capacity of the second capacitor is set to be smallerthan, or equal to the electrostatic capacity of the first capacitor,then it is possible to suppress that the charging voltage of the firstcapacitor becomes low. As a result, while increasing of the turn-OFFloss is suppressed, the surge voltage can be suppressed.

In accordance with a fourth modification mode of the present disclosedideas, a switching circuit is arranged by a transistor, a capacitor, anda diode having an anode terminal and a cathode terminal. While thetransistor has a control electrode, a first electrode, and a secondelectrode, the transistor controls a first electrode current flowingthrough the first electrode based upon a signal entered to the controlelectrode. The first electrode of the transistor is connected via thecapacitor to the anode terminal of the diode. The control electrode ofthe transistor has been connected to the cathode terminal of the diode.The control electrode is either a gate or a base. The first electrode iseither a drain or a collector.

In accordance with the above-described circuit, in the switching circuitcontaining the transistor, the surge voltage can be suppressed in acorrect and stable manner.

Alternatively, as an alternative idea, in the case that a voltage at thefirst electrode is 0 V, a capacitance between the control electrode andthe first electrode is defined as a “control-to-first electrodecapacitance”, and the capacitance may alternatively have such acapacitance which is larger than the control-to-first electrodecapacitance by 0.01 time to 100 times.

Alternatively, as an alternative idea, the diode may have a pconductivity type anode semiconductor region and an n conductivity typecathode semiconductor region. Both the anode semiconductor region andthe cathode semiconductor region have been embedded in a front surfaceof a semiconductor substrate.

Alternatively, as an alternative idea, the diode may have a pconductivity type anode semiconductor region and an n conductivity typecathode semiconductor region. The anode semiconductor region has beenprovided via an insulating film on the semiconductor substrate. Whilethe cathode semi conductor region has been provided via an insulatingfilm on the semiconductor substrate, this cathode semiconductor regionhas been contacted to the anode semiconductor region. The capacitor hasan embedded conductive region and a coating insulation region. Theembedded conductive region is elongated from a first plane to a secondplane of the semiconductor substrate. The coating insulation region hascoated the embedded conductive region in such a manner that the embeddedconductive region is electrically insulated from the semiconductorsubstrate. The cathode semiconductor region has been electricallyconnected to the control electrode of the transistor. The anodesemiconductor region has been connected to the embedded conductiveregion. The embedded conductive region has been electrically insulatedvia the coating insulation region from the first electrode of thetransistor.

Alternatively, as an alternative idea, the diode may have a p wellregion, an anode semiconductor region, and a cathode semiconductorregion. The p well region is expanded from the first plane to the secondplane of the semiconductor substrate, and has a p conductivity type. Theanode semiconductor region is arranged on a surface layer of the p wellregion of the semiconductor substrate, and has the p conductivity type.The cathode semiconductor region is arranged on the surface layer of thep well region of the semiconductor substrate, and has the n conductivitytype. The capacitor has an embedded conductive region and a coatinginsulation region. The embedded conductive region is elongated from thefirst plane to the second plane of the semiconductor substrate. Thecoating insulation region has coated the embedded conductive region insuch a manner that the embedded conductive region is electricallyinsulated from the semiconductor substrate. The cathode semiconductorregion has been electrically connected to the control electrode of thetransistor. The anode semiconductor region has been connected to theembedded conductive region. The embedded conductive region has beenelectrically insulated via the coating insulation region from the firstelectrode of the transistor.

In accordance with a fifth modification mode of the present disclosedideas, a switching circuit is arranged by a transistor, a capacitor, anda diode having an anode terminal and a cathode terminal. While thetransistor has a control electrode, a first electrode, and a secondelectrode, the transistor controls a first electrode current flowingthrough the first electrode based upon a signal entered to the controlelectrode. The first electrode of the transistor is connected to theanode terminal of the diode. The control electrode of the transistor hasbeen connected via the capacitor to the cathode terminal of the diode.The control electrode is either a gate or a base. The first electrode iseither a drain or a collector.

In accordance with the above-described circuit, in the switching circuitcontaining the transistor, the surge voltage can be suppressed in acorrect and stable manner.

Alternatively, as an alternative idea, the diode may have a p wellregion, an anode semiconductor region, and a cathode semiconductorregion. The p well region is expanded from a first plane to a secondplane of a semiconductor substrate, and has a p conductivity type. Theanode semiconductor region is arranged on a surface layer of the p wellregion of the semiconductor substrate, and has the p conductivity type.The cathode semiconductor region is arranged on the surface layer of thep well region of the semiconductor substrate, and has the n conductivitytype. The capacitor has an embedded conductive region and a coatinginsulation region. The embedded conductive region is elongated from thefirst plane to the second plane of the semiconductor substrate. Thecoating insulation region has coated the embedded conductive region insuch a manner that the embedded conductive region is electricallyinsulated from the semiconductor substrate. The embedded conductiveregion has been electrically connected to the control electrode of thetransistor. The cathode semiconductor region has been electricallyinsulated via the coating insulation region from the embedded conductiveregion. The anode semiconductor region has been electrically connectedto the first electrode of the transistor.

Alternatively, as an alternative idea, the diode may have a p wellregion, an anode semiconductor region, and a cathode semiconductorregion. The p well region is expanded from the first plane to the secondplane of the semiconductor substrate, and has a p conductivity type. Theanode semiconductor region is arranged on a surface layer of the p wellregion of the semiconductor substrate, and has the p conductivity type.The cathode semiconductor region is arranged on the surface layer of thep well region of the semiconductor substrate, and has the n conductivitytype. The capacitor has an embedded conductive region and a coatinginsulation region. The embedded conductive region is elongated from thefirst plane to the second plane of the semiconductor substrate. Thecoating insulation region has coated the embedded conductive region insuch a manner that the embedded conductive region is electricallyinsulated from the semiconductor substrate. The region has beenelectrically connected to the control electrode of the transistor. Thecathode semiconductor region has been electrically insulated via thecoating insulation region from the control electrode of the transistor.The cathode semiconductor region has been electrically connected to theembedded conductive region. The anode semiconductor region has beenelectrically connected to the first electrode of the transistor.

In accordance with a sixth modification mode of the present disclosedideas, a driving circuit for driving a transistor is arranged by avariable resistor. The transistor has a control electrode, a firstelectrode, and a second electrode. The variable resistor has beenelectrically connected to the control electrode of the transistor. Thevariable resistor has a depletion layer which is expanded/compressed inresponse to a voltage between the first electrode and the secondelectrode of the transistor. The depletion layer can control a width ofa current path of the variable resistor.

The above-described driving circuit does not utilize ON/OFF operationsof a semiconductor switching element. The driving circuit utilizes thedepletion layer which is expanded/compressed in response to a voltagebetween the major electrodes of the transistor so as to adjust the widthof the current path of the movable resistor. When the depletion layer isexpanded so that the width of the current path of the variable resistorbecomes narrow, the resistance value of the variable resistor isadjusted to be large. When the depletion layer is compressed so that thewidth of the current path of the variable resistor becomes wide, theresistance value of the variable resistor is adjusted to be small. Thedepletion layer is expanded/compressed in response to the voltagebetween the major electrodes of the transistor, and this event iscontinuous with respect to increase/decrease of the voltage between themajor electrodes of the transistor. A circuit used to correctly set athreshold value is not required, which is however required for ON/OFFoperations of a semiconductor switching element. As a consequence, thearrangement of the above-described driving circuit is made simple.

The above-descried driving circuit utilizes the depletion layer which isexpanded/compressed in response to the voltage between the majorelectrodes of the transistor so as to adjust the width of the currentpath of the variable resistor. Also, the arrangement of the drivingcircuit can be made simple, and therefore, manufacturing cost thereofcan be suppressed to low cost.

Alternatively, as an alternative idea, the variable resistor may have avariable resistance value. In the case that the voltage between thefirst electrode and the second electrode of the transistor is low, thevariable resistance value is adjusted to be small, whereas in the casethat the voltage between the first electrode and the second electrode ofthe transistor is high, the variable resistance value is adjusted to belarge. In accordance with this modification mode, in a beginning stageof a transition time period during which the transistor is turned OFF, aresistance value of a gate resistor is adjusted to be small, whereas ina final stage of the transition time period, the resistance value of thegate resistor is adjust to be large. As a consequence, in the beginningstage of the transition time period for turning OFF the transistor, theresistance value of the gate resistor can be adjusted to be small, sothat the gate current can be steeply varied. As a result, the draincurrent of the transistor is steeply varied, so that the time requiredfor turning OFF the transistor can be shorted. Furthermore, in the finalstage of the transition time period for turning OFF the transistor, theresistance value of the gate resistor can be adjusted to be large, sothat the gate current can be gently varied. As a result, the draincurrent of the transistor is gently varied, so that the increasing ofthe surge voltage can be suppressed. In accordance with the drivingcircuit of this modification mode, the characteristic as to thetransition time period for turning OFF the transistor can be improved.

Alternatively, as an alternative case, the variable resistor may have ap type semiconductor region, and one pair of n type semiconductorregions. One pair of the n type semiconductor region is contacted to thep type semiconductor region, and sandwich this p type semiconductorregion. The p type semiconductor region has electrically connected tothe control electrode of the transistor. One pair of the n typesemiconductor regions have been electrically connected to the firstelectrode of the transistor. In this variable resistor, the p typesemiconductor region corresponds to a current path. If the voltagebetween the major electrodes of the transistor becomes high, then a pnjunction between the p type semiconductor region and the n typesemiconductor regions is reverse-biased, so that the depletion layer isexpanded to the p type semiconductor region. As a consequence, when thevoltage between the major electrodes of the transistor becomes high, thewidth of the current path is adjusted to become wide. On the other hand,if the voltage between the major electrodes of the transistor becomeslow, the depletion layer which has been expanded to the p typesemiconductor region is compressed. As a consequence, when the voltagebetween the major electrodes of the transistor becomes low, the width ofthe current path is adjusted to become wide. Accordingly, thecharacteristic as to the transition time period for turning OFF thetransistor can be improved.

Alternatively, as an alternative idea, the driving circuit mayfurthermore contain a zener diode. The zener diode has been arrangedbetween one pair of the n type semiconductor regions and the firstelectrode of the transistor. If the zener diode is employed, then thevoltage between the major electrodes of the transistor is not applied tothe n type semiconductor regions of the variable resistor until thezener diode breaks down. As a consequence, until the zener diode breaksdown, the width of the current path of the variable resistor is keptwide. As a result, in the beginning stage of the transition time periodduring which the transistor is turned OFF, while the resistance value ofthe variable resistor is kept small, the gate current can be steeplyvaried. As a result, the drain current of the transistor can be steeplyvaried, so that the time required to turn OFF the transistor can befurthermore shortened. It should also be noted that a plurality of zenerdiodes may be alternatively series-connected to each other. Since theplurality of zener diodes are employed, the voltage to be applied to then type semiconductor regions may be adjusted.

Alternatively, as an alternative idea, the driving circuit may furtherhave a first diode, a second diode, a first resistor, and a drivingvoltage generating circuit. The variable resistor has been electricallyconnected via the second diode to the driving voltage generatingcircuit. The driving voltage generating circuit has been further andelectrically connected via the first resistor and the first diode to thecontrol electrode of the transistor.

Alternatively, as an alternative case, the variable resistor may have ap type semiconductor region, one pair of insulation regions, and onepair of conductive regions. One pair of the conductive regions iscontacted to the p type semiconductor region, and sandwich this p typesemiconductor region. One-paired conductive regions are located to the ptype semiconductor region via one-paired insulation regions. The p typesemiconductor region has been electrically connected to the controlelectrode of the transistor. One pair of the conductive regions havebeen electrically connected to the first electrode of the transistor. Inthis case, the p type semiconductor region has been electricallyconnected to the gate electrode of the transistor. The conductive regionhas been electrically connected to the output electrode of thetransistor. The p type semiconductor region, the insulation regions, andthe conductive regions constitute an MIS (Metal Insulator Semiconductor)structure. In the variable resistor, the p type semiconductor region isa current path. When the voltage between the major electrode of thetransistor becomes high, the depletion layer is expanded to the p typesemiconductor region due to the electric field effect of the MISstructure. As a consequence, when the voltage between the majorelectrodes of the transistor becomes high, the width of the current pathis adjusted to become narrow. On the other hand, if the voltage betweenthe major electrodes of the transistor becomes low, then the depletionlayer which has been expanded to the p type semiconductor region iscompressed. As a consequence, when the voltage between the majorelectrodes of the transistor becomes low, the width of the current pathis adjusted to become wide. In the driving circuit containing theabove-described variable resistor, the characteristic as to thetransition time period during which the transistor is turned OFF can beimproved.

Alternatively, as an alternative idea, both the transistor and thevariable resistor may be formed in the same semiconductor substrate. Ifthe transistor and the variable resistor have been formed in the samesemiconductor substrate, then the transistor and the variable resistorneed not be prepared as separate circuit components. If the transistorand the variable resistor have been formed in the same semiconductorsubstrate, then the driving circuit may be arranged by employing a smallnumber of circuit components, and furthermore, the driving circuit maybe made compact with practical better utilization.

Alternatively, as an alternative case, the variable resistor may have ap type semiconductor region, and one pair of n type semiconductorregions. One pair of the n type semiconductor regions are contacted tothe p type semiconductor region, and sandwich this p type semiconductorregion. The p type semiconductor region has electrically connected tothe control electrode of the transistor. The p type semiconductor regionhas been arranged via the insulating film on the semiconductorsubstrate. One pair of the n type semiconductor regions have beenelectrically connected to the first electrode of the transistor. Onepair of n type semiconductor regions have been arranged via theinsulating film on the semiconductor substrate.

Alternatively, as an alternative idea, both the p type semiconductorregion and one pair of the n type semiconductor regions may be made ofpoly crystalline silicon. If poly crystalline silicon is employed, thenboth the p type semiconductor region and the n type semiconductor regionmay be alternatively and readily formed on the semiconductor substrateby utilizing a manufacturing process of a semiconductor device.

In accordance with a seventh modification mode of the present disclosedideas, a driving circuit for driving a transistor is arranged by anadjusting circuit. The transistor has a control electrode, an inputelectrode, and an output electrode. The adjusting circuit adjusts aresistance value of the control electrode of the transistor based upon acontrol current flowing through the control electrode of the transistor.

The Inventors of the present disclosure have paid their attentions togate currents of the transistor. In a transition time period for turningON the transistor and another transition time period for turning OFF thetransistor, magnitudes of gate currents of this transistor are changedin a time elapse manner. As a consequence, if a magnitude of a gatecurrent value of the transistor is employed as a parameter, then acondition under which the transistor is operated can be monitored. Ifthe magnitude of the gate current value of the transistor is employed asthe parameter, then a resistance value of a gate resistor can beadjusted while the adjustment of the gate resistance value is turned tothe operation of the transistor.

In other words, the above-described driving circuit is featured byemploying such an adjusting circuit capable of adjusting the resistancevalue of the gate resistor of the transistor based upon the currentvalue of the gate current of the transistor. Also, the driving circuitadjusts the resistance value of the gate resistor of the transistorbased upon the gate current which is capable of reflecting operationstates of the transistor under better condition. As a result, thedriving circuit can adjust the resistance value of the gate resistorwhile being turned to the operations of the transistor.

Alternatively, as an alternative idea, the adjusting circuit may adjusta resistance value of the control electrode of the transistor based upona preset threshold value of a control current.

In the above-described adjusting circuit, a predetermined current valueof the gate current has been set as the threshold value. The adjustingcircuit adjusts the resistance value of the gate resistor based uponthis threshold value. As a consequence, a driving circuit containing theabove-explained adjusting circuit can drive the transistor by correctlydiscriminating a beginning stage of a transition time period for turningOFF the transistor from a final stage thereof.

Alternatively, as an alternative idea, the adjusting circuit may adjusta resistance value of the control electrode of the transistor based upona control current when electron charges are discharged which have beenstored in the control electrode of the transistor. A driving circuitcontaining this adjusting circuit drives a field-effect type transistor.In accordance with the adjusting circuit, while being turned to such anoperation when the transistor is turned OFF, the resistance value of thegate resistor of the transistor can be adjusted. The characteristic whenthe transistor is turned OFF can be improved.

Alternatively, as an alternative idea, the driving circuit may furthercontain a driving voltage generating circuit. The driving voltagegenerating circuit has been electrically connected via a fixed resistorand a diode to the control electrode of the transistor. The diode has ananode and a cathode. The anode has been connected via the fixed resistorto the driving voltage generating circuit. The cathode has beenconnected to the control electrode of the transistor. An ON-voltagesignal generated from the driving voltage generating circuit isconverted into a gate current by the fixed resistor, and electroncharges are supplied to the gate electrode of the transistor. As aresult, the driving circuit transfers the operation state of thetransistor to an ON state. Furthermore, when the ON state of thetransistor is transferred to an OFF state, the electron charges whichhave been stored in the gate electrode of the transistor can flow as anegative gate current to the adjusting circuit, since a signal pathflowing through the fixed resistor is cut off by the diode. Theadjusting circuit can adjust the resistance value of the gate resistorof the transistor based upon this negative gate current.

Alternatively, as an alternative idea, the diode, the adjusting circuit,and the transistor may be arranged within the same semiconductorsubstrate.

Alternatively, as an alternative idea, when an absolute value of acontrol current is large, the adjusting circuit may decrease theresistance value of the control electrode, whereas when an absolutevalue of a control current is small, the adjusting circuit may increasethe resistance value of the control electrode. If this adjusting circuitis provide, then within the transition time period during which thetransistor is turned OFF, in the beginning stage where the absolutevalue of the gate current value becomes large, the adjusting circuit candecrease the gate resistor so that the electron charges can be quicklydischarged. On the other hands, in the final stage where the absolutevalue of the gate current value becomes small, the adjusting circuit canincrease the gate resistor so that the electron charges can be gentlydischarged. As a consequence, in the above-described driving circuit, inthe beginning stage of the transition time period for turning OFF thetransistor, the drain current flowing through the transistor is steeplyvaried, so that the time required for turning OFF the transistor can beshortened. Moreover, in the final stage of the transition time periodfor turning OFF the transistor, the drain current flowing through thetransistor is gently varied, so that increasing of the surge voltage canbe suppressed. According to the above-described driving circuit, thetrade-off relationship between the surge voltage and the turn-off losscan be overcome.

Alternatively, as an alternative case, the adjusting circuit may containan adjusting-purpose transistor and a resistor. The resistor has alarger resistance value than an ON resistance of the adjusting-purposetransistor. The adjusting-purpose transistor has a control electrode, aninput electrode, and an output electrode. The resistor has been arrangedbetween the control electrode and the input electrode of theadjusting-purpose transistor. The resistor has been connected to thecontrol electrode of the transistor. The adjusting-purpose transistor isturned ON/OFF in response to a voltage difference generated between bothterminals of the resistor. When an absolute value of a control currentof the transistor is large, the adjusting-purpose transistor is turnedON, so that the control current of the transistor passes through theadjusting-purpose transistor. When an absolute value of a controlcurrent of the transistor is small, the adjusting-purpose transistor isturned OFF, so that the control current of the transistor passes throughthe resistor. In accordance with the above-described driving circuit,within the transition time period for turning OFF the transistor, in thebeginning stage where the absolute value of the gate current valuebecomes large, the adjusting-purpose transistor is turned ON based uponthe voltage difference generated between the both terminals of theresistor, so that the gate current flows through the adjusting-purposetransistor. Since the ON resistance of the adjusting-purpose transistoris small, the electron charges which have been stored in the gateelectrode of the transistor can be quickly discharged. On the otherhand, in the final stage where the absolute value of the gate currentvalue becomes small, since the voltage difference generated between theboth terminals of the resistor become low, the adjusting-purposetransistor is turned OFF, so that the gate current flows through theresistor. Since the resistance value of the resistor is large, theelectron charges which have been stored in the gate electrode of thetransistor can be gently discharged. As a result, in accordance with theabove-described driving circuit, the trade-off relationship between thesurge voltage and the turn-OFF loss can be overcome.

Alternatively, as another alternative idea, the adjusting-purposetransistor may be an p type field-effect transistor. The input electrodeof the adjusting-purpose transistor has been connected between theresistor and the control electrode of the transistor. The outputelectrode of the adjusting-purpose transistor has been grounded.

Alternatively, as further alternative idea, the adjusting-purposetransistor may be a n type field-effect transistor. Both the controlelectrode and the output electrode of the adjusting-purpose transistorhave been connected between the resistor and the control electrode ofthe transistor.

While the invention has been described with reference to preferredembodiments thereof, it is to be understood that the invention is notlimited to the preferred embodiments and constructions. The invention isintended to cover various modification and equivalent arrangements. Inaddition, while the various combinations and configurations, which arepreferred, other combinations and configurations, including more, lessor only a single element, are also within the spirit and scope of theinvention.

What is claimed is:
 1. A driving circuit for driving a transistorcomprising: a control circuit, wherein the transistor includes a controlelectrode, an output electrode and an input electrode, the controlcircuit controls a control electrode resistance of the transistor basedon a control current flowing through the control electrode of thetransistor, the control circuit controls the control electroderesistance of the transistor based on a predetermined threshold of thecurrent, the control circuit controls the control electrode resistanceof the transistor based on the control current when a charge accumulatedin the control electrode of the transistor is being discharge, thecontrol circuit controls the control electrode resistance to decreasewhen an absolute value of the control current is large, the controlcircuit controls the control electrode resistance to increase when theabsolute value of the control current is small, the control circuitincludes an adjustment transistor and a resistor, the resistor has aresistance which is larger than an on-state resistance of the adjustmenttransistor, the adjustment transistor includes a control electrode, anoutput electrode and an input electrode, the resistor is disposedbetween the control electrode and the input electrode of the adjustmenttransistor, the resistor is coupled with the control electrode of thetransistor, the adjustment transistor is turned on and off in accordancewith a voltage difference generated between both ends of the resistor,the adjustment transistor is turned on so that the control current ofthe transistor flows through the adjustment transistor when an absolutevalue of a control current of the transistor is large, the adjustmenttransistor is turned off so that the control current of the transistorflows through the resistor when the absolute value of the controlcurrent of the transistor is small, the adjustment transistor is a Pconductive type field effect transistor, the input electrode of theadjustment transistor is coupled between the resistor and the controlelectrode of the transistor, and the output electrode of the adjustmenttransistor is grounded.
 2. The driving circuit according to claim 1,further comprising: a driving voltage generation circuit, wherein thedriving voltage generation circuit is electrically coupled with thecontrol electrode of the transistor through a fixed resistor and a diodetherebetween, the diode includes an anode and a cathode, the anode iscoupled with the driving voltage generation circuit through the fixedresistor therebetween, and the cathode is coupled with the controlelectrode of the transistor.
 3. The driving circuit according to claim2, wherein the diode, the control circuit and the transistor aredisposed in a same semiconductor substrate.
 4. A driving circuit fordriving a transistor comprising: a control circuit, wherein thetransistor includes a control electrode, an output electrode and aninput electrode, the control circuit controls a control electroderesistance of the transistor based on a control current flowing throughthe control electrode of the transistor, the control circuit controlsthe control electrode resistance of the transistor based on apredetermined threshold of the control current, the control circuitcontrols the control electrode resistance of the transistor based on theconcurrent when a charge accumulated in the control electrode of thetransistor is being discharged, the control circuit controls the controlelectrode resistance to decrease when an absolute value of the controlcurrent is large, the control circuit controls the control electroderesistance to increase when the absolute value of the control current issmall, the control circuit includes an adjustment transistor resistor,the resistor has a resistance which is larger than an on-stateresistance of the adjustment transistor, the adjustment transistorincludes a control electrode, an output electrode and an input electrodethe resistor is disposed between the control electrode and the inputelectrode of the adjustment transistor, the resistor is coupled with thecontrol electrode of the transistor, the adjustment transistor is turnedon and off in accordance with a voltage difference generated betweenboth ends of the resistor, the adjustment transistor is turned on sothat the control current of the transistor flows through the adjustmenttransistor when an absolute value of a control current of the transistoris large, the adjustment transistor is turned off so that the controlcurrent of the transistor flows through the resistor when the absolutevalue of the control current of the transistor is small, the adjustmenttransistor is an N conductive type field effect transistor, the controlelectrode and the output electrode of the adjustment transistor arecoupled between the resistor and the control electrode of thetransistor.
 5. The driving circuit according to claim 4, furthercomprising: a driving voltage generation circuit, wherein the drivingvoltage generation circuit is electrically coupled with the controlelectrode of the transistor through a fixed resistor and a diodetherebetween, the diode includes an anode and a cathode, the anode iscoupled with the driving voltage generation circuit through the fixedresistor therebetween, and the cathode is coupled with the controlelectrode of the transistor.
 6. The driving circuit according to claim5, wherein the diode, the control circuit and the transistor aredisposed in a same semiconductor substrate.
 7. A driving circuit fordriving a transistor comprising: a control circuit, and a drivingvoltage generation circuit, wherein the transistor includes a controlelectrode, an output electrode and an input electrode, wherein thecontrol circuit controls a control electrode resistance of thetransistor based on a control current flowing through the controlelectrode of the transistor, wherein the control circuit controls thecontrol electrode resistance of the transistor based on the controlcurrent when a charge accumulated in the control electrode of thetransistor is being discharged, wherein the driving voltage generationcircuit is electrically coupled with the control electrode of thetransistor through a fixed resistor and a diode therebetween, whereinthe diode includes an anode and a cathode, wherein the anode is coupledwith the driving voltage generation circuit through the fixed resistortherebetween, wherein the cathode is coupled with the control electrodeof the transistor, and wherein the control circuit controls the controlelectrode resistance to be reversely proportional to an absolute valueof the control current.